1 electrical measurement of a high frequency, high
TRANSCRIPT
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ELECTRICAL MEASUREMENT
OF A HIGH FREQUENCY, HIGH CAPACITANCE
PIEZOCERAMIC RESONATOR WITH RESISTIVE ELECTRODES
Alex V. Mezheritsky, “ELECTRICAL MEASUREMENT OF A HIGH FREQUENCY,
HIGH CAPACITANCE PIEZOCERAMIC RESONATOR WITH RESISTIVE ELECTRODES”,
adapted from the article published in the journal IEEE UFFC, Jan. 2005.
ABSTRACT - In a thin and large area PZT-ceramics piezoresonator (PR) with relatively low resonance
impedance, caused by high frequency resonance and high PR capacitance, the effect of electrode resistivity
and parasitic resistive and inductive elements in the measurement fixture results in significant distortion of the
measured thickness-mode (longitudinal TL, shear TS) resonance response - resonance frequency shifts and
characteristics deformation. This distortion may not allow the precise measurement of the PR characteristic
frequencies, quality factor and electro-mechanical coupling coefficient so essential to a complete PR and
material characterization. A theoretical description of the “energy-trap” phenomena in a thickness-vibrating PR
with resistive electrodes is presented. To interpret electrical measurements, the electro-mechanical model,
including for completeness both the PR with resistive electrodes (as a system with “distributed parameters”)
and the measurement fixture, is developed. The method of two contact points on the electrode provides deep
sharpening and exact determination of the PR resonance. For the optimal disposition of the contact fingers the
resonance bandwidth of a real PR with resistance electrodes is even more pointed than that for the ideal PR.
Keywords : piezoceramics, resonator, resonance frequency, electrode, fixture, energy trap.
Author: A.V. Mezheritsky e- mail: [email protected] t ( CC: [email protected])
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I. INTRODUCTION
Thin piezoplates made from piezoelectric ceramics are widely used as high-frequency resonators,
transducers, and monolithic filters employing thickness modes of vibration. The plate is polarized and covered
with metallic electrodes in order to excite it with an applied RF field by means of the piezoelectric effect.
The complete characterization of piezoelectric thickness-mode resonators [1] includes measurements of the
parameters which indicate the effectiveness of the electromechanical energy conversion. Such a basic parameter,
the coefficient k of electro-mechanical coupling (CEMC), can be defined as the difference between the material
elastic stiffness i jcE,D [2],[3] at constant strength E and induction D of an electric field. In an elementary theory,
the thickness-mode CEMC k is practically determined from resonance frequency data of PR impedance Z [2],
neglecting other modes influence:
( )2
0
tan 211
2KH
Z ki C KHω
= −
, (1)
where 0C Hε= ΦS is the PR capacitance, 2 2k h cε= S D is the CEMC squared, 2 2K cω ρ= D is the
thickness mode wavenumber squared, h is the piezocoefficient, ε S is the dielectric permittivity, H is the PR
thickness, Φ is the electrode area, ρ is the piezomaterial density. The piezomaterial constants [2] set
, , ,c h kε S D for the TS vibrational mode is 11 44 15 15, , ,c h kε S D , and for TL mode is 33 33 33, , , tc h kε S D ,
respectively. The typical frequency responses of a hypothetical PR are shown in Fig. 1. Such characteristics have
both the resonance rf and antiresonance af frequencies [3], defined here as frequency extremes of the output
voltage (absolute value). The IEC Standard [2] determines the CEMC squared according to a practical expression
for the fundamental thickness harmonic as 21 0.405 0.81rk δ= + , where 1r a rf fδ = − is the relative resonant
frequency interval.
In general, thickness vibrations are very complicated, since coupling of different modes may occur in the
vicinity of the fundamental resonance [4]-[7]. So, the basic TL-mode with a coupled TS vibration, or the basic
TS-mode with coupled flexure-twist vibrations, under certain conditions, can provide the “energy-trapping”
effect widely used in frequency selection devices. Application of such simple expressions to piezoelectric
3 ceramic materials is difficult due to the influence of competing modes of vibration and the effects that are
caused by boundary conditions. Typically, these result in response distortions leading to erroneous
characteristic frequency values. Several techniques have been developed to reduce the difficulties that result
from these effects.
Fig 1 (a,b,c). Comparative frequency responses (c) with the resonance rf and antiresonance af frequencies
of a thickness-mode PR for traditional (a) and proposed (“new”) (b) measurement techniques with in series
connected parasitic inductance L and resistor R in the PR fixture.
One of them [8] determines the value of k based on the shift in location of the overtone resonance
frequencies. This technique is well suited for overcoming the problems associated with the use of rδ
determining. The overtone frequencies of the thickness-mode are sufficiently high [1] to reduce the problems
associated with any competing lateral modes. Also, for a thickness-mode PR the influence of the boundary
effects is less for overtones than it is for the fundamental mode. However, the distortions of the frequency
responses of a high capacitance PR at high frequencies are effected in a manner that is not a result of unwanted
mode or boundary effects. Rather, this behavior is related, as to material parameters, to the high dielectric
constant ( 0i jε εS ; 100… 5000, where 0ε is the vacuum dielectric permittivity) of the piezoceramic (PZT)
materials [9],[10]. This so-called “inversion response” effect was firstly reported in [11], and then was
described in details in [12],[13]. The observed resonance frequency shift and inversion effect for higher
harmonics are the result of the relationship between the specimen capacitance and the holding fixture series
4 inductance. The essential shift and distortion of the resonance peak (Fig. 1a) of a system including PR and an
electrical circuit is determined by inductive and resistive elements in the part of the circuit common for the PR
input and output circuitry loops.
Modern practical needs of PR applications at high frequencies 10…30 MHz, and higher, require usage of thin
plates of 200…50 mc thickness or even less. Meantime, large area piezotransducers (up to tens sq. cm) have
some applications to produce a wide wave-front radiating effect (in medicine, etc.), or are used for technology
optimization (as intermediate large area piezoelements for tiny delay line piezotransducers, etc.). All these lead
to high PR capacitance (tens nF), and hence, to low resonance PR resistance (much less 1 Ohm) and low
equivalent PR inductance (tens nH), so the very small inductive and resistive parasitic elements in the
measurement fixture and PR electrode resistance have an increased significance. The thinner piezoplate
thickness, the thinner electrode thickness is required. Particularly for this reason, limited methods can be
applied for electrode deposition. It also leads to worsening of electrical properties of a thin electrode layer [14]
which has lower conductivity than pure metal. The same situation concerns thin and thick piezo-films [15],
and bulk multilayer piezotransducers [16] consisting of multiple thin piezolayers co-fired through thin metal
layers. Determination of such PRs resonance properties is a problem.
II. ELECTR0-MECHANICAL MODEL DESCRIPTION
A. Basic Estimations
The IEC and IEEE Standards [2],[3] were used as a guideline to conduct the frequency measurements for
this study. The circuit with a “grounded” PR shown in Fig. 1 (a regime of a constant current through PR
( )EI E R const f≅ =% ) was utilized [17] in order to make the measurements. Two designs for the specimen
holding fixture were modeled - traditional “one contact point” and proposed new “two contact points”
measurement techniques. A TS-mode PR type was chosen for experiments as less acoustically sensitive to
lateral boundaries and planar spurious resonances. The PR 20x10x0.2 mm was made from PZT-36 ([9],
Russia) with polarization direction along a 10 mm side, the resonance fr = 4.0 MHz and antiresonance
fa = 5.4 MHz frequencies (CEMC 15k = 0.7, rδ = 0.28), electric capacitance C0 = 10 nF, mechanical quality
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factor ( ) 100mQ Q = . As the intrinsic resonance and antiresonance PR resistances (ideal electrodes) are:
2 2 208r rR C k Qπ ν ω≅ , 2 2 2 2
08 (1 )a aR k Q C kπ ν ω≅ − , (2)
where , ,2r a r afω π= , we have a numerical value estimation for the fundamental harmonic ( 1ν = ):
0.1rR Ohm; , 242aR Ohm; . The fully metallized PR has chemically deposited Ni - electrodes 20x10 mm
with a thickness of 2 mc and electric resistance elR of 3 Ohm between the farthest points on the electrode,
so that the estimated ratios 2 0.13r elR R ; and 2 0.8el rR R Ohm; .
The best results for a linear PR amplitude-frequency characteristics (AFCh) performance were obtained when
using relatively large PR input electrical load ER = 1.5 kOhm >> aR (Fig. 1). From the magnitude display,
the antiresonance frequency can be discerned with ease. The resonance frequency, however, cannot be
determined accurately by traditional inspection. For estimating the influences of the decisive factors of
the fixture wire inductance and resistance, and resistance of the PR electrodes, it is needed to take into account
two basic intrinsic PR parameters of the resonance PR resistance rR and equivalent inductance 1L :
2
20
1( )r
i j
HR
k Qν
ε ε⋅ ⋅
Φ S∼ ,
3
1 20
1( )
( )i j
HL const
kν
ε ε⋅ =
Φ S∼ , (3)
then 2 21 01 0.32rL C k Hω µ≅; . The series inductance in the metal finger that holds the test specimens in
place and its resistance are comparable to the intrinsic PR resonance resistance rR and equivalent inductance
L1 values. By an approximate estimation, a contact wire of 1 cm length with parasitic inductance 0.1L Hµ;
and typical quality factor 10Q ; has effective resistance 0.1R Ohm; at 5 MHz according to R L Qω≈ .
A PR that displayed normal responses when operating at their fundamental frequency also displayed inverted
responses at their overtone frequencies and their shifts are not be related directly to CEMC. If some inductance
L is connected to PR in series, there is a shift νω∆ of the resonance frequency of ν -harmonic
, 10.5 ( )r L L constν νω ω ν∆ − =; . The antiresonance frequency is not affected very much by the series
inductance, while the resonance frequency is lowered. Since the frequency distance between two nearby
harmonics is twice of fundamental resonance , 12 rω; , the “inversion response” effect definitely occurs when
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the frequency shift , 1rνω ω∆ ≥ takes place at the harmonics ( , ,1r rνω ν ω⋅; ) of the following order
3
12
0
2 1 1( )i j
L HL L k
νε ε
≥ ⋅ ⋅Φ S
∼ (4)
The level of the effect increases to the degree that the “resonance frequency” of the higher harmonic appears on
the positive side of the antiresonance peak of the lower frequency harmonic . Note that two times decreased PR
thickness (characteristic frequencies) causes an increase in critical harmonic number of the inversion response
effect by an order.
The presence of fixture wire and PR electrode resistance increases a voltage response mostly at the resonance
frequency and, hence, leads to a resonance bandwidth widening. It can even result in an absence of any well-
defined resonance frequency peak in the expected frequency range. In the case of resistive PR electrodes (non-
equipotential electrode surface), an effectively excited PR region decreases significantly, so only resonance
characteristics of a local PR region can be detected. A new “two contact points” measurement technique was
proposed and analyzed, which fully eliminate described above effects and, even more, transforms a negative
feature into positive one as to electrode resistivity. In the present discussion, the electrodes are assumed to be so
thin that their mechanical effects can be neglected.
B. Introductory Description of the Electrode Resistivity Effect on “Uncoupled”, Infinite Piezoplate
A linear one-dimensional semi-infinite piezoplate (strip) and its equivalent representation (Fig. 2, regions
A-B) are analyzed. Elementary space units are considered here acoustically independent, or “uncoupled”,
in the lateral x-direction. A direct practical analog of such a model can be a 3-1 or 3-2 piezocomposit.
The coupling-effect influence for a regular PR is considered in the next chapter.
An exciting input voltage inU is applied to the strip-points 0 0′− and an output voltage outU is picked-off
from the strip-points n n′− on the electrodes. Such a PR represents a system with “distributed” parameters
[18],[19] consisting of infinitely repeated r z r− − chains with elr R x= ∆ and 11 z Z x−= ∆ , where elR and
1Z − are the linear electrode resistance and PR intrinsic bulk admittance (1) (per unit length), respectively.
According to [18], the effective input PR complex impedance inZ is determined by ( ) 11 1 2in inZ z r Z−− −= + + ,
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then 2 2 2in el el inZ R Z R Z x= − ∆ , and at 0x∆ → we finally have :
2 2 ( ) ( )in el elZ R Z R t Z t= = , (5)
where ( )elR t , Z(t) are the electrode resistance between points x = 0, t and the intrinsic impedance of the PR
region (0, )x t∈ , respectively, t is an arbitrary chosen point on the electrode. According to the equivalent
schema representation of Fig. 2, the voltage distribution is determined as ( )( ) 1 2n
out in inU U x U r Z≡ = + .
Fig. 2. PR frequency response measuring circuit and its equivalent representation for a semi-infinite piezoplate.
As 0r → at 0x∆ → , we have in the limit of n → ∞ (n x x∆ = ) the exponential voltage decay:
( ) exp( 2 ) exp( ) exp( )in el in RU x U R x Z x i K xµ= − ⋅ = − = − ⋅ , (6)
in inU I Z=
where the basic distribution parameters 2R elK i R Z= − as an analog of the wavenumber, and
1 2R eli K Z Rµ = = as a parameter of relative electrode conductivity, are used further in acoustical and
electrical approaches. If RK is expressed as Re ImR R RK K i K= + , then the absolute value of the voltage
( ) exp(Im ) exp( )in R RU x U K x x x= ⋅ ≡ − , (7)
where 2
1 Im ReR Rx K µ µ= − = is the characteristic dimension of an effectively excited PR region for
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Im 0RK < at 0x > , that corresponds to an e-times voltage magnitude decay along the PR electroded surface.
Further, as the PR admittance can be represented as 11 1 1Re Im iZ Z i Z Z e φ−− − −= + = , where φ is the phase
of the intrinsic PR admittance, then 11 Im cos(0.5 ) 2R R elx K R Zφ
−− = − = ⋅ . For pure capacitive admittance
10Z i Cω− = with the linear PR capacitance 0C (per unit length), we have the decay parameter
01C R elCx x R Cω≡ = , (8)
At the PR resonance with the intrins ic resonance resistance rR (per unit length) we have
20 , ,2 ( ) 2 ( ) 2R r r el r el C rx x R R t R t R t x k Qν≡ = = ≅ ⋅ . (9)
According to (5, 9), an estimation of ,in rZ and 0x at the resonance frequency of a real finite PR can be made
as follows: , 2in r el rZ R R; and 0 2r elx M R R; , where rR is the intrinsic PR resonance resistance
estimated according to (2) using piezoceramic material parameters, elR is the measured electrode resistance
between the farthest points on the electrode, M is the lateral PR dimension. Initial estimation gives
0 2.6x mm; and , 0.8in rZ Ohm; for the experimental specimen.
Taking into account that ( )EI E R const f≅ =% and in inU I Z= , we have the expression for the output voltage
magnitude outU picked-off from the contact separated with the input contact by distance W :
( )1
( ) exp 2 exp cos(0.5 ) 2out in R el elx WU U x I Z W x I R Z R Z Wφ
−
=
≡ = − = ⋅ − ⋅ ⋅
. (10)
Since the antiresonance PR resistance is greater as much as orders the resonance one, minimal size of an
effectively excited PR region corresponds exactly to the resonance thickness frequency. In particular, for a real
PR with finite dimensions at least near the PR antiresonance with el aR R<< we have Rx → ∞ ( Rx M>> ,
where M is the PR length), and then inZ Z= , out inU U= for any x.
Note that the presented linear description can be easily extended to a PR radial configuration with contact
dot-points on the PR electrodes.
9 III. ELECTRODE RESISTIVITY INFLUENCE ON “ENERGY - TRAP” EFFECT
The vibrations of acoustically coupled space units generally lead to a vibrational modes coupling which is
primarily determined by a lateral gradient of the basic mode vibrational characteristics. In the classical “energy-
trap” phenomenon [5], the gradient is created by sufficiently different electrical boundary conditions on the
electroded (short-circuited) and unelectroded (open-circuited) plate surfaces. So, in a TL-mode PR, a
concomitant coupled (TS) mode is mostly localized in a narrow boundary between the electroded (excited)
and unelectroded (unexcited) regions [4],[5].
In the case of resistive electrodes the gradient is wider distributed in the lateral directions and hence is less
localized. In the limit conditions of infinite (ideal) and zero electrode conductivity, the results of the two
approaches trend to equal dispersion relationships. Moreover, if the piezomaterial parameters of elastic
anisotropy ( 33 44 4E Ec c > ) provide sufficiently strong support of the “energy-trap” effect, the only electroded
region in a partly electroded piezoplate vibrates. Then, the piezoplate can be hard-fixed in the surrounding
unelectroded region without influence on the vibrations of the PR active zone, which can be considered as
independent of surroundings. Such an effect is fundamental in high-frequency resonator and monolithic filter
designs based on “energy trapping”. For this reason, it appears to expect that in certain conditions for a PR with
resistive electrodes the vibrational regions can be considered as acoustically uncoupled because of relatively
small planar gradient. Further the theory of the “energy-trap” effect in a TL-mode PR with resistive electrodes
is developed based on the approach presented in [4], whose mathematical procedures and original notations are
generally used.
A. Common Description
A piezoceramic plate polarized and excited in its thickness direction can vibrate in TL as well as TS modes
which are coupled. The dispersion spectrum of a circular disk PR is derived in [4] from the constitutive
equations. The problem is reduced to a two-dimensional one according to the full rotational symmetry of a disk
PR. All variables are radial and axial components only. A trapped energy PR of this kind consists of a thin
ceramic disk with an inner electroded area where propagation of TL-modes is possible, surrounded by a
nonelectroded marginal region where the modes decay outwards. Allowance is made for different electrical
10 boundary conditions such that electroded and nonelectroded regions of the disk may be distinguished. Thus if
energy trapping is to work, the appropriate dispersion curves must allow, at the resonance frequency, a real
planar wavenumber xK in the electroded region to correspond with an imaginary xK in the unelectroded
region. This is only true for frequencies within a certain respective "energy trapping” interval.
If, however, an interchange of modes occurs, there is no trapping interval for the TL-mode and thus energy
trapping does not work.
The faces of the piezoplate are located at 2z H= ± . The rotation axis of the disk coincides with the z-axis,
which is also the direction of polarization. The electric field is determined by its electric potential as
( ) ( )z x z xE ϕ ′= − . A harmonic time dependence exp( t)iω is assumed for all vibration characteristics.
In order to solve the equations of state simultaneously the variables are separated by
( , ) ( ) ( )zu x z F x f z= ⋅ , ( , ) ( ) ( )x xu x z F x h z′= ⋅ , 33( , ) ( ) ( ) ( )x z F x g z h f zϕ = ⋅ − ⋅ , (11)
where ( )z xu are the local displacements, ( ) , ( ), ( )f z g z h z are some z-coordinate functions. Inserting this into
equations of charge and motion yields the differential equation for radial distribution function ( )F x as
2 0x x x xF F x K F′′ ′+ + = , where xK is the separation constant and turns out to be the appropriate “planar”
wavenumber. As can be shown, a similar procedure can be applied to a finite thin piezoplate equally excited
along an infinite line , e.g. 0x = for certainty. Solving the task leads to (11) with the lateral equation
2( ) ( ) 0x x xF x K F x′′ + ⋅ = , (12)
which will be considered in further analysis. For a disk PR the radial dependence as a solution of Bessel’s
differential equation is simply given by zero order Bessel functions 0( )xK x⋅¢ , for a linear PR configuration
the solution of (12) is in the exponential form exp( )xi K x⋅ . The axial variations, however, must be found from
a system of three coupled equations. To discover their exact solutions represents a fairly complicated problem
which, however, can be simplified considerably by imposing the only restrictive condition between
wavenumbers - the planar wavenumber xK is taken to be small compared with the “axial” quasi-wavenumbers
of the TL and TS modes. In this way the analysis is confined to the fundamental thickness vibrations of a thin
11 plate with planar dimensions much larger its thickness.
For z-variables there are ultimately three basic boundary equations to be solved. Two of them are related to
the mechanical boundary conditions - for thickness vibrations no stresses are allowed to be exerted at the major
faces of the plate. They must be traction-free, so that the stresses z zT and x zT must vanish at 2z H= ± ,
where the linear piezoelectric constitutive relations are used.
The traditional electrical boundary conditions on the plate surfaces are a pair of shorted and grounded
electrodes covering the major PR surfaces for which ( )2
0z H
xϕ=±
= (called e-region), or a pair of
unelectroded and charge-free surfaces - no current is allowed to pass the surfaces of the plate - for which
( )2
0z z HD x
=±= (s-region). For an unelectroded surface, the effect of the electric field in the surrounding
space can be considered by requiring that both the electrical potential and the normal component of the
dielectric displacement vector cross the interface continuously.
In the present discussion, the electrodes are assumed to be so thin that their mechanical effects can be
neglected. The above equations and boundary conditions are homogeneous, and constitute an eigenvalue
problem. Both the boundary conditions applied to the local variations of displacements and potential yield
the dispersion relations in both the electroded e-region and the nonelectroded s-region of the plate.
The dimensionless frequency 440.5 EH cω ρΩ = , related to the second TS resonance, is used for the
dispersion relations. Solving the differential equations together, we have the matrix of coefficients in a system
of linear equations for vibrational amplitudes [4]:
, , ,11 12 13
21 22 23
31 32 33
e s e s e sa a aA a a a
a a a= , (13)
with the requirement 0A = for a non-trivial solution. The sets of 2ia and 3ia are initially appear from
the two mechanical boundary conditions, the last set ,1e s
ia corresponds to “electroded” (e – index) and
‘unelectroded” (s – index) major plate surfaces. Then, ( )xK Ω can be found, providing zero-order Bessel
functions 0 ( )xK x⋅¢ for radial symmetry, or exponential functions exp( )xK x⋅ for a linear task with
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the same A-matrix. Usually, Ω is plotted versus the planar propagation constant xK , which is then found to
be an ambiguous function split up into several branches. Each branch represents the dispersion relationship
between Ω and xK for a definite mode of propagation. A real value of xK means propagation of that
respective mode with a planar linear dependence ( )exp Re xi K x⋅ , whereas an imaginary xK indicates
a mode that decays in 0x > direction exponentially with ( )exp Im xK x⋅ for Im 0xK < .
Fig. 3 illustrates the dispersion curves for PCM A [4] with relatively la rge “energy trap” figure-of-merit
33 44 4c c >E E . The TL mode lies well above the thickness-radial-shear mode (TS) in both the electroded e-region
and the unelectroded s-region. The two modes are connected with imaginary continuation loops which,
however, close at values 2 0.4xK H > beyond the scope covered by the approximation made. As was shown
[4], small deviations in material parameters cause a conversion of the spectrum with an apparent interchange
of modes - which differ slightly in their material properties but differ largely in their dispersion. This has its
marked influence on the phase and group velocities of the respective waves and on the frequencies of the
thickness vibrational modes.
B. Piezoplate with Resistive Electrodes
In the case of resistive electrodes each elementary unit volume is acoustically coupled and electrically loaded
by surrounding regions. If alternative voltage 2
0.5 inz HUϕ
=±= ± is applied to some point on the linear PR
electrodes, then
( ) ( )2 t
2
1x zz H x
z Hel
x D x dxh
ϕσ
∞
=±=±
′′ ± = − ∫ , or ( ) ( )2 2
1xx zz H z H
el
x i D xh
ϕ ωσ=± =±
′′± = , (14)
where σ is the unit electrode conductivity, elh is the electrode thickness. It means that for an elementary
electrode unit with length dx the difference of the currents on its left and right sides equals the surface normal
inductance. Then, following the procedures of [4], the equation (14) can be transformed into
( )2 11 33 1( ) 2 0e s
x i el iK H a i H h aωσ ε−⋅ + ⋅ =S , (15)
and then the A-matrix is modified for the case of resistive electrodes as follows:
13
2 2 211 11 12 12 13 13
21 22 23
31 32 33
( ) ( ) ( )e s e s e sx x xK H a G a K H a G a K H a G a
A a a aa a a
⋅ + ⋅ ⋅ + ⋅ ⋅ + ⋅= , (16)
where ( )1 233 02 2el elG i H h i C R Hωσ ε ω−= = ⋅S . From the condition 0A = for non-trivial solutions the
value of ( )xK Ω can be found. As (16) is a superposition of the e- and s-cases, generally there are two sets
of ( )xK Ω roots depending on electrode resistivity and corresponding to the respective dispersion curve.
Fig. 3. A common view of the dispersion curve ( )2xK H Ω of coupled TL-TS vibrational modes of a PR
with resistive electrodes for the case of 33 44 4E Ec c > ( PCM A [4], 100Q = ): - level of a given frequency, - line of constant electrode conductivity,
- dispersion dependence on electrode conductivity at a given frequency, .
- energy-trap dispersion curves for an electrodless and fully electroded (infinite conductivity)
piezoplate.
Using , 0( ) 1C E C E el Ex x R Cω ω ω≡ = = as an intermediate parameter, where 33E
E rc Hω π ρ ω= ;
(short- circuited electrodes generally does not mean local 0=E ), then
14
2
,33 44
4E E
C E
HG i
xc cπ
Ω=
. (17)
The dispersion relationships ( )xK Ω for a thin plate with resistive electrodes were calculated and plotted (Fig.
3) as a space graph in perpendicular ( )Re 2xK H and ( )Im 2xK H planes for PCM A [4] with basic material
parameters: 33 0 190Sε ε = , 11 0 425Sε ε = , 0.51tk = , 15 0.65k = , 33 44 4.55c c =E E , 10 233 19.4 10Dc N m= ⋅ ,
10 244 5.5 10Dc N m= ⋅ , and then comparisons were conducted for PCMs which differ in their parameters [9].
Internal energy dissipation was taken into account by representing elastic constants ( )ˆ 1i j i jc c i Q= +E E as
complex, where Q is the intrinsic material (resonance) quality factor. Calculated for the traditional lossless
Q = ∞ and lossy 100Q = cases, the basic dispersion e- and s-curves have similar character, but in a lossy
case the dependence generally is not purely real or imaginary, so that it never intersects the Ω -axis near PR’s
resonance and antiresonance. The dispersion ( )Re 2xK H vs. ( )Im 2xK H dependences on electrode
conductivity (as a relative ,Cx E parameter) for a lossy and lossless PCM are shown in Fig. 4 a,b for
antiresonance (a) and resonance (b) frequency regions, represented as frequency “cuts” of Fig. 3.
For the antiresonance region the resistance-dependent dispersion lines go out from the s-branch
corresponding to the extremely high electrode resistivity elR → ∞ , and then come into the Ω-axis with
0xK H → for relatively high electrode conductivity 0elR → . In particular, above the PR antiresonance,
where the PR admittance is predominantly capacitive, ( )Im 2xK H− has a maximum corresponding to the
condition 0 1elC Rω ; of maximum energy dissipation on a resistive electrode. In general, there are two sets of
A-matrix roots for a given frequency, corresponding to the respective branch. In the PR resonance interval near
the resonance frequency, the first root-set going out from the s-branch provides ( )exp 1xK x → , as 0xK H →
at 0elR → , and the second root-set coming into the e-branch is an effective non-zero solution. As was
established by simulations, for relatively high electrode conductivity with , 1CH x <<E , when
15
1xK H Q<< , the ( )Im 2xK H of the first root-set is sufficiently linear in respect to the resistive
parameter ,CH x E (Fig. 5), so that this case is under further detailed analysis. The second solution showing
( )Im 2 1xK H Q− > corresponds to a rapidly decaying distribution function with characteristic decay
parameter comparable with the plate thickness. As seen from Fig. 4b, there is a point of singularity near the PR
resonance on the dispersion surfaces (Fig. 3), which is a basic factor of maximum dispersion distortion and
non-linearity.
Fig. 4 a,b. Dispersion curves of TL-TS coupled modes for a PR with resistive electrodes near the PR
antiresonance (a) and resonance (b) frequencies ( Ω values shown): - equal-resistance lines,
- internal lossy case with 100Q = , - internal lossless case with Q = ∞ ,
- limit e- and s-branches with internal losses, - the point of singularity. PCM A [4].
A comparison of two, “uncoupled” and “coupled”, approaches is presented in Fig. 5. Dependences of the
direct decay parameters ( )( )Im 0.5 x RK H− for the both cases on the relative parameter of electrode resistivity
, 1CH x <<E were calculated for several symmetric characteristic frequencies near the piezoplate TL-resonance
(AFCh for a “uncoupled” case is represented inside the figure). First dependence (dotted lines) is based on
expression (7) for an “uncoupled” case and shows a linear character. Second dependence (solid lines) derived
on the base of expression (16) corresponds to a “coupled” case and shows strong non-linearity mostly into a
16 very narrow 3 dB frequency bandwidth near basic TL-resonance. The two dependences coincide for relatively
low electrode resistivity, when the two approaches, “uncoupled” and “coupled”, give similar results for the
( )( )Im 0.5 x RK H decay parameters. At the TL-resonance ( )2( ) ,0.5Im 0.5x R C Er
K H k Q H x− → ⋅ , that
corresponds to the maximum slope of the dispersion curves. Based on dispersion simulations on wide-range
material parameters [9], the following relationship for the point of singularity (Fig. 4b) was empirically
established
( ) 0
0.5Im min 2x x RK H K H Qβ
∗ →− ≅; , (18)
where 2
2 233
44
2 3(1 ) 14 tc
kc
β
− ≈
E
E; is the parameter of piezo-elastic anisotropy influence. Applying a 0.5
coefficient for the 10% difference between Im xK and Im RK , we have a linearity threshold condition at the
TL resonance
0 1Im x r
QxH K H β
= − > (19)
Fig. 5. Dependence of the imaginary part of the planar wavenumber on electrode resistivity near the PR TL-
resonance for PCM A [4]. Solid lines are for Im xK from “energy-trap” equations, and dotted lines are for
Im RK from “uncoupled” approach – all for 100Q = . Dashed lines are for the resonance with Q = ∞ .
17
Note that the Im xK non-linearity, taking place mostly at the TL resonance, is a piezomaterial dissipative effect
low-sensitive to the figure-of-merit 33 44c cE E , so critical for classical “energy trapping”. The “weak electrode
resistivity” approach with characteristic decay distance more as much as an order of the plate thickness is
considered further, providing a simple engineering description of the measurement technique mostly important
for practice.
IV. “WEAK ELECTRODE RESISTIVITY” APPROACH
A. Effective Quality Factor of an Infinite Piezoplate with Resistive Electrodes
Let us further carry out a comparative analysis of precision features for the two systems of resonance
frequency measurements based ultimately on the frequency dependences ( )inU f and ( )outU f for a given
electrode resistivity of an infinite piezoplate under the condition (19) 10Q H x Mβ − < << . The intrinsic PR
admittance (ideal electrodes) in a vicinity of the fundamental resonance peak with 2 1k Q >> is described [20]
by ( ) (1 )rZ f R i y+; , then ( ) 21rZ f R y+; , where 2y Q χ= is the generalized resonance frequency
displacement, 1rf fχ = − is the relative frequency displacement, Q is the intrinsic material quality factor.
The sharpness of a resonance peak is defined by the frequency bandwidth at a level of 3 dB, or 2 , of the
resonance factor magnitude. For the ideal PR out inU U Z= ∼ , then 2 1 2(1 )outU y+∼ , and the PR resonance
quality factor 3r r dBQ f f Q= ∆ = .
In the case of resistive electrodes, for the traditional schema with a one-point contact (Fig. 1a) in the regime
of a frequency-constant current, according to (6): out in inU U I Z Z= = ∼ , then 2 1 4(1 )outU y+∼ . Then,
the PR input quality factor equals :
3 3r r dBQ f f Q= ∆ =% , (20)
that means a ~1.7 wider resonance peak.
For the proposed schema with two-point contacts (Fig. 1b) separated by distance W, in the regime of
a frequency-constant current, the output voltage frequency dependence near the resonance is determined in
18 accordance with (6,7) as:
( )( ) ( ) 2 1 4
2 1 40
cos(0.5arctan )(1 ) exp 1
(1 )out
out r
U f W yW y
U f x y
≅ + ⋅ − + , (21)
Fig. 6. Effective quality factor dependence vs. 0W x parameter at the resonance frequency for two-point
contacts on the resistive electrode separated by distance W. The solid line is predicted from (23), and the solid
dots are experimental data for the specimen with 0 2.6x mm= and 20M mm= .
Then, we can define the effective PR output quality factor ( )effQ W as a measure of output resonance peak
sharpness as follows:
( ) 3eff r dBQ W f f Q N= ∆ = , (22)
where the N-factor is determined by
( )
[ ]
2
02 1 4
0.5 ln2 0.25 l n 1
cos 0.5 arctan( )1
(1 )
NWNx
N
− +≅
−+
(23)
As seen from Fig. 6, when the distance between contacts 0W x> , the output resonance bandwidth of a PR with
resistive electrodes is more pointed typically as much as 2 – 3 times even of the case of the ideal PR. For a two-
sided infinite piezoplate (“A” plus “B” regions with “C” region on the left, Fig. 2), the exponential voltage
distribution factor does not change, but the effective PR impedance inZ , and hence outU , becomes twice lower.
19 B. Finite Piezoplate Frequency Responses (Electrical Lateral Boundary Effect)
The basic dimension parameters of a finite PR with resistive electrodes are presented in Fig. 7. Lateral PR
dimensions are supposed to be relatively large in respect to the PR thickness, so that the “uncoupled” approach
can be further used (19) for 0 Qx HM Mβ
> ⋅ , or as an estimation for a finite plate 2
22r
el
R H QR M β
> ⋅
.
From the continuity of currents along the electrode, the basic equation for voltage distribution
( ) ( ) ( )2 2z H z H
U x x xϕ ϕ= =−
= − is expressed as:
12 ( )m
x el xU R Z U x dx−′− = ∫ , (24)
which can be further transformed into the differential equation to be solved 2 ( ) 0xxU U xµ ′′ − = , where
2 elZ Rµ = is the voltage distribution parameter, with the common solution in the form
1 2( ) exp( ) exp( )U x P x P xµ µ= − + , where 1 2,P P are coefficients. Applying the electrical boundary
conditions (0) inU U= and 2
0 0( )
m
x xU U x dxµ
→+′− = ∫ , the total current through PR is expressed as:
( ) ( )010 0( ) 0
( ) ( )m
x x elx xM mI Z U x dx U x dx U U R−
→− →+− −′ ′= + = −∫ ∫ . (25)
Finally we have the expressions for the output voltage and effective input PR admittance:
( ) ( , )out inU W W m U= ℑ ⋅ , in inU I Z= ⋅ , ( )1 1,inZ M m Zµ− −= Ψ ⋅ , (26)
exp( ) exp( )( , )
1 exp( 2 ) 1 exp(2 )W WW m
m mµ µ
µ µ−ℑ = +
+ − + , (27)
( ) ( ) ( ) ( ) ( )1 1 1 1,
1 exp 2 1 exp 2 1 exp 2 1 exp 2M m
m m M m M mµ µ µ µ
Ψ = − + − + − + + − − + −
, (28)
where ( , )W mℑ is the voltage transfer function, ( ),M mΨ is the coefficient of the boundary effect influence,
m is the distance between the input contact point and plate corner in the direction of the output contact point on
the electroded PR surface, so that W < m < M , [0;1]W M ∈ , [0;1]m M ∈ ,
1[ ; ]Rx M Q H Mβ −∈ ∞ (7).
20
Fig. 7 (a,b). Basic geometry of a real finite PR plate (a) with resistive electrodes and input and output contacts
(the shadow density reflects the electric field intensity distribution), and lateral voltage distribution (b) along the
electrodes.
Once the PR voltage transfer function is known, it can be incorporated into the electrical measurement
system. Different particular variants of PR connection with corresponding basic expressions are summarized in
Appendix, Fig. 11 (A1-A5). The terms “center” and “corner” mean that the point of the applied input voltage is
located at the plate center with 2m M= , or at the plate corner with m M= , respectively, with “flowing”
output contact.
The traditional technique (Fig. 11 A1, A3) with one contact point is described by (26-28) with
( , ) 1 W mℑ = ( W = 0 ). For this case, comparative AFCh for connected in series inductance L and resistor R
are presented in Fig. 8a for the PR 20x10x0.2 mm with electrical load 1.5ER kOhm= . The influence of the
parasitic fixture elements result in resonance peak shift and distortion - it becomes hardly detectable.
The effect of L and R influence strongly depends on the point of the contact.
The new technique (Fig. 11 A2) with two spaced contact points is described by (26-28). Comparative AFCh
for both techniques are presented in Fig. 8b. For the new technique (Fig. 1b) with two spaced contacts there is
no influence on the resonance response from fixture parasitic inductance and resistance elements connected at
any place in the schema practically up to at least 10L Hµ= and 10R Ohm= , several orders greater than the
21 corresponding PR equivalent parameters. It takes place due to the absence of the part of the circuitry common
for the PR input and output loops. The resonance peak becomes sharp and detectable with ease. Experimental
AFCh fully correspond to those simulated with the “uncoupled” approach (Fig. 8) , as to resonance peak shape,
amplitude, and frequency shifts.
Fig. 8 (a,b). Simulated AFCh for the traditional schema A3 (Appendix) (a) with fixture inductance L and
resistance R for different locations of the contacts on the plate, and input and output AFCh for schemas A1, A2
(Appendix) (b) with different connections. PR has two resistive electrodes with 3elR Ohm= , M = 20 mm, W =
20(10) mm. Dashed line is for intrinsic PR AFCh ( 0elR = ).
Generalized dependences of relative PR resonance responses on the parameter 0x M of electrode
conductivity in comparison to the ideal PR ( outU I Z= with 0elR = ) and their distribution along the PR
electroded surface are shown in Fig. 9 for a finite PR plate. There is a maximum of output voltage outU
magnitude under 0x (electrode conductivity) variation at the PR resonance corresponding to 0x W= for
0.4W M< in a “corner” configuration. The resonance peak amplitude for a real PR is less than that for the
ideal PR when the condition 0.4W M> is satisfied [21] in a wide range of electrode conductivity. Note that
the input voltage 0in out W
U U=
= amplitude at the PR resonance is inversely proportional to the product of
effective quality factor (22) and capacitance of the effectively excited PR zone near the input point.
22
Fig. 9. Relative PR responses vs. parameter 0x M of electrode conductivity at the resonance frequency for a
set of input-output distances W ( ( )I const f= ). Finite PR plate.
Since the voltage distribution factor for a finite PR in comparison to the case of infinite PR (6) can be
expressed as ( )exp( ) ,out inU U W W mµ= − ⋅ ℜ , then
( , ) exp( ) ( , )W m W W mµℜ = ⋅ ℑ , (29)
and similar for the output voltage ( )2 exp( ) , ,out elU I R Z W Dim W M mµ= − ⋅ with
( ), , exp( ) ( , ) ( , )Dim W M m W W m M mµ= ⋅ ℑ Ψ , (30)
where ( , )W mℜ and ( ), ,Dim W M m are the boundary “correction” coefficients to the case of a finite plate.
Taking place an increase of the output response (resonance peak widening) is caused by the influence of the
boundary effect (Fig. 10). The two cases of finite and infinite plates can be considered approximately equal for
the condition 0 0.5x M< and 02m x> at the resonance under “uncoupled” approach restrictions.
In the limit of relatively high electrode conductivity with 1Rx M << it follows from (26-28) that:
inZ Z µ= and ( ) exp( )W W µℑ = − , then ( )expoutU I Z W µ µ= − when the input contact is placed at
23
the corner of the plate, and 2inZ Z µ= and ( ) exp( )W W µℑ = − , then ( )exp 2outU I Z W µ µ= −
when the input contact is placed at the center of the plate. These results are similar to the case of infinite PR.
Fig. 10. Distribution of relative, in respect to the ideal PR, resonance output responses along the PR electrode
for different points ( x ) the input voltage applied to, Wx x W− = , 0x M = 0.26. Solid lines are predicted from
(26-28), and the solid dots are experimental data for the specimen 20x10x0.2 mm.
V. CONCLUSION
In thin (200…50 mc) and large area (tens sq. cm) PZT-ceramics PRs with low resonance impedance (much
less 1 Ohm), caused by high frequency (tens MHz) and high PR capacitance (tens nF), the effect of electrode
resistivity and parasitic resistive and inductive elements in the measurement fixture results in significant
distortion of the measured thickness-mode resonance frequency response, including high-order PR harmonics.
The theory of the coupled-modes “energy-trap” effect in a PR with resistive electrodes was developed.
As the effect of electrode resistivity influence is dissipative, the lateral gradient of vibrational characteristic s,
being proportional to Q , allows to apply an “uncoupled” approach to the PR resonance behavior description
for typical piezoceramic materials.
24 A simple fixture principle with two contact points on the electrode was proposed providing deep sharpening
and exact determination of the resonance frequencies even in the presence of the parasitic factors due to absence
of common with the PR input and output circuitry loops. As a result, for optimal disposition of the contact
fingers on the finite piezoplate the resonance bandwidth of a real PR with resistive electrodes is more pointed
than even in the case of the ideal PR. There is no need any additional compensative elements or to make very
accurate measurements of the parasitic elements that are associated with a specimen holding fixture to
algebraically subtract their influence.
The proposed analysis and method can be also applied to bulk multilayer piezo-transducers consisting of
multiple thin piezo-layers co-fired through thin metal layers, and can be used particularly for investigating of
resonance properties distribution (inhomogeneity) along the surface of a thin piezoplate (film).
REFERENCES
[1] T. R. Meeker, "Thickness mode piezoelectric transducers,” Ultrasonics, vol. 10-1, pp. 26-36, 1972.
[2] IEC Standard. Guide to Dynamic Measurements of Piezoelectric Ceramics with High Electromechanical
Coupling, Publication 483, 1976.
[3] IRE Standards on Piezoelectric Crystals - The Piezoelectric Vibrator: Definitions and Methods of
Measurement, Proc. IRE, vol. 45, no. 3, pp. 353-358, 1957.
[4] P. Schnabel, “Dispersion of thickness vibration of piezoceramic disk resonators,” IEEE Trans. Ultrason.,
Ferroelect., Freq. Contr., vol. SU-25, no. 1, pp. 16-24, 1978.
[5] R. Holland and E.P. EerNisse, Design of resonant piezoelectric devices, Cambridge: MIT Press, 1968.
[6] A.A. Comparini and J.J. Hannon, “Flexural, Width-Shear, and Width-Twist vibrations of thin rectangular
crystal plates,” IEEE Trans. Ultrason., Ferroelect., Freq. Contr., vol. SU-21, no. 2, pp. 130-135, 1974.
[7] D.C. Gazis and R.D. Mindlin, “Extensional vibrations and waves in a circular disk and a semi-infinite
plate,” J. App. Mech., vol. 27, pp. 541-547, 1960.
25 [8] M. Onoe, H.F. Tiersten, and A.H. Meitzler, "Shift in the location of resonant frequencies caused by large
electromechanical coupling in thickness-mode resonators," JASA, vol. 35, no. 1, pp. 36-42, 1963.
[9] OCT 11 0444-87. Piezoceramic Materials. Specifications, Moscow: ElectronStandard, 1988.
[10] D.A. Berlincourt, D.R. Curran, and H. Jaffe, "Piezoelectric and piezomagnetic materials and their function
in transducers", Physical Acoustics, vol. 1, part A, Editor - W.P.Mason, New York: Academic Press, 1964.
[11] A.V. Mezheritsky, Theoretical and Experimental Investigation of Complex Electro-Elastic Constants of
Piezoceramics, Moscow Institute of Physics and Technology (MIPT), Ph.D. Dissertation, 1985.
[12] S.N. Wickstrom and T.R. Meeker, "Electrical measurement of high frequency ceramic resonators,"
Proc. IEEE Ultrasonics Symposium, pp. 717 – 722, 1989.
[13] A.V. Mezheritsky, “Measurement of resonance characteristics of high frequency piezoceramic
resonators with high electric capacitance,” Measurement techniques (Russian), no. 9, pp. 53 – 55, 1990.
[14] S. Sherrit, N. Gauthier, E.D. Wiederick, B.K. Mukherjee, and S.E. Prasad, “The effect electrode
materials on measured piezoelectric properties of ceramics and ceramic-polymer composites,”
Proc. IEEE Ultrasonics Symposium, pp. 346 – 349, 1991.
[15] R.A. Dorey and R.W. Whatmore, “Apparent reduction in the value of the d33 piezoelectric coefficient in
PZT thick films,” Integrated Ferroelectrics, vol. 50, pp. 111 – 119, 2002
[16] Y. Sasaki, S. Takahashi, and M. Yamamoto, “Mechanical quality factor of multilayer piezoelectric
ceramic transducers,” Jpn. J. Appl. Phys., vol. 40, part 1, no. 5B, pp. 3549-3551, 2001.
[17] G.L. Ragen, Microwave Transmission Circuits, New York: Dover Publications, 1965.
[18] R.P. Feynman, R.B. Leighton, M.L. Sands, The Feynman Lectures on Physics. Electrodynamics,
Massachusets: Addison Wesley, 1963.
[19] O.P. Gribovsky, Ceramic Solid Circuits, Moscow: Energy, 1971.
[20] A.V. Mezheritsky, “Quality factor of piezoceramics,” Ferroelectrics, vol. 266, pp. 277-304, 2002.
[21] A.V. Mezheritsky, “Method of measurement of the resonance frequencies of piezoceramic resonator,”
Russia patent 1273840, Patent Bulletin no. 44, 1986
26
APPENDIX
There are a number of different variants of PR connection, which are presented in Fig. 11 (A1-A5).
Traditional schema
with parasitic elements in the input circuitry loop only.
out in
E in
U ZE R i L R Zω
=+ + +% A 1 )
( , )out in
E in
U Z D mE R i L R Zω
⋅ ℑ=
+ + +% A 2 )
Traditional schema with parasitic elements in the part of the circuit common for input and output loops.
out in
E in
U Z i L RE R i L R Z
ωω+ +
=+ + +% A 3 )
0.5 1 ( , )inout
E in
Z D m i L RUE R i L R Z
ωω
+ ℑ + +=
+ + +% A 4 )
(both resistive electrodes)
( , )out in
E in
U Z D m i L RE R i L R Z
ωω
⋅ ℑ + +=
+ + +% A 5 )
(the only upper resistive electrode)
Fig. 11 (A1-A5). Variants of PR connection and corresponding basic expressions for PR responses.
An increased number of electrical contacts in the proposed PR fixing system definitely increases probability
of a contact failure in one of them. Such situations were modeled for the researched PR 20x10x0.2 mm (M = 20
mm) with input electric load 1.5ER kOhm= (Fig. 12).
27
Fig. 12 (a,b,c). Simulated AFCh for schema A4 (Appendix) under variation of:
a - inductance L (at R = 0) for W = 8 mm, b - inductance L (at R = 0) for W = 20 mm,
c - resistance R (at L = 0) for W = 20 and 10 mm. Dashed line is for intrinsic PR AFCh ( 0elR = ).
The schema A4 (Appendix) with “accidentally” disconnected output “ground” contact was considered. In this
case the effect of the fixture parasitic inductance and resistance takes place again, and the character of the
influence depends greatly on the point of the output contact. Characteristic features of the dependences are
connected to a specific voltage phase rotation along the electrode, and further voltages superposit ion.
Experimental AFCh, corresponding to schema A4 conditions, show satisfactory coincidence with those
presented in Fig. 12, as to resonance peak shape and frequency shifts.
28 Glossary
( ), , ( ) ,r r aZ Z R R Rφ – PT impedance, its phase, resonance and antiresonance resistances (linear values)
,r z , inZ – elementary electrode resistance and PR impedance in the model, effective input PR impedance
i jcE,D, i jh , i jε S
, i jk (k ) – piezomaterial constants (generalized notation)
, , ,D E T S – electric field induction and strength, mechanical stress and strain
, ( ) ,m r effQ Q Q Q – standardized PCM , PR resonance, and effective quality factors
0 0 1( ),C C L – PT capacitance (linear value), and its equivalent inductance
, ,f ω Ω – frequency, angular and dimensionless frequencies
, , ,r a rf f δ ν – PR resonance and antiresonance frequencies, relative resonance frequency interval, and overtone
number , yχ – frequency displacement and generalized frequency displacement
Eω – TL-mode characteristic frequency ( E rω ω; )
,x z – space coordinates
( ), ,el el elR R h σ – electrode resistance (linear value), its thickness and unit resistance
, ,in outU Uϕ – electric potential, PT input and output voltages
, , , ,EE I R R L% – schema electromotive force, total current through PR, load, parasitic resistance and
inductance
K and ,x RK K – complex thickness and lateral wavenumbers
0, , ,R Cx x x µ – characteristic decay parameters
, ,M H Φ – PR length, thickness, and electrode area
, , , ,WW m x x t – parameters of PR measurement configuration
, , , ,Dim Fℑ Ψ ℜ – voltage distribution parameters ,, ,e s
i jA a G – matrix, its elements, and parameter of electrode resistivity influence
,x zu u – local PT displacements
( ) , ( ) , ( )f z g z h z – z-coordinate vibrational functions
29
30
The author A.V. Mezheritsky : 525 Ocean Pkwy, # 3 J, Brooklyn, NY 11218 [email protected] Ph.D. in Physics (1985, MIPT, Moscow, Russia), IEEE Member