Å > ¦ × ç

Ú ] ˙ û ˝ F

î = d

䟚£>2ämU¹äÂDŸd¾Î¹äÂ

UWB Subharmonic Quadrature-IF Mixer and Image

Rejection Mixer

û ˝ Þ : ’Ì

N û ` ¤ : "/± ²=

2M¬Å þ ý ~

䟚£>2ämU¹äÂDŸd¾Î¹äÂ

UWB Subharmonic Quadrature-IF Mixer and Image

Rejection Mixer

û ˝ Þ : ’Ì Student : Chih-Hao Yang

N û ` ¤ : "/± ²= Advisor : Dr. Chi-Yang Chang

Å > ¦ × ç

Ú ] ˙ ç Í

î = d

A ThesisSubmitted to Institude of Communication EngineeringCollege of Electrical Engineering and Computer Science

National Chiao Tung Universityin Partial Fulfillment of the Requirements

for the Degree ofMaster of Science

inCommunication Engineering

June 2008HsinChu, Taiwan, Republic of China

2M¬Åþý~

䟚£>2ämU¹äÂDŸd¾Î¹äÂ

û˝Þ: ’Ì N û ` ¤: "/± ²=

Å >¦×çÚ]˙û˝F

¿ b

Ì(¦m2í äÍ$, MÚƉAÑø_ÏÆ1/½bí¦mÍ$, äÍ$íÔ4Ê

kË'×íä £0§íf§0 ¾Î äÍ$¹äÂ52íŸdmU, uTôÍ$4

?í^j¶ 7Ê äÍ$¹äÂç2, ä90¯Âu|½bíjK

…d퇚¨, BbUàÖ-ZVõÛø_ äí90¯Â, w2¯#

|#í¶M, ‚àò¯ÂV®A ?Óª¯Â¥¦¸D½"4^?í^k-Z6

ʤ¹d2n

…dí(š¨Üs¹ä øus_|Î5È90óPÏퟚ

£>2ämU¹äÂ, ù†ux¾ÎŸdmUŠ?ퟚ¹ä œüíÞUà

œýíjKb, u¥s¹äÂíiõ

i

ii

UWB Subharmonic Quadrature-IF Mixer and Image RejectionMixer

Student: Chih-Hao Yang Advisor: Dr. Chi-Yang Chang

Institude of Communication EngineeringNational Chiao Tung University

Abstract

The UWB system is receiving growing attention as an important communication

system for wireless communication. The characteristics of UWB system are the

ultra wide bandwidth and its high data rate transmission. In order to obtain a

better performance of the UWB system, an UWB mixer with the function of re-

jecting image signal is needed. An ultra broadband directional coupler is the most

important element of the UWB mixer due to the large bandwidth.

In first part of this thesis, we will utilize multisection structure to perform an

UWB coupler. The most tightly coupling section of the multisection coupler is

fabricated by VIP structure. The compensated stub, which can improve return loss

and isolation of this coupler, is presented in this thesis.

In the second part of this thesis, two kinds of mixers are discussed. The first

one is a subharmonic quadrature-IF mixer, the phase difference between two output

ports is 90 degree. The second one is a subharmonic image rejection mixer. All the

mixers mentioned above have the benefits of small layout area and fewer elements.

iii

iv

Acknowledgement

Ðá

ÊBíÞ·2rÖA, Fbí|Û, UBí°ç5˜ß¯rÖ, 1)J¤êABíî=

d

íl, Bb>áBíNû`¤ "/±²=, ÊFNûísu~2, .cÇó7BçXû

˝í ‰e…, "`¤f~í`çG, ¢bÊó¤çÞ®šíø…, ?UDÞ¿>â61J

5Ñ\š `¤JAâÇ[º.[óíj÷=Bíû˝G, ,u_v/ŸË#8‡D

Nû, É-çÞ1q™TËÑçÞO; "`¤6ÊBböÅív`, .‘<íBb

®Þº%ð, 1/É-BbÊ“5ÕíÞº ÊBí-ñ2, "`¤uøP,uÞ

®ñ, .éçÞ>§ƒ9F>íç6, ?DÊ`¤íNû5-dû˝, Bg)í•/

«

wŸ, >á¨tãºn`¤ 0`¤ Ï£2`¤, ú¤dT|^~í<c, F

bíNû, U…d?›ê¾

y6, >áõðf~íçÅÏN ÛïS ‰ †> D£Y ÏÀÃ, ÊB`

Xƒ@½v, Fb,uÏ-í6ŒB, TXB.ý<c£õ¶, öí'ááFb °¢íÏn

D ÅAý ŠT lT ˜.d, 'òEû˝Fsí¬˙2, ?DFbAóT, D

Fbø–Uı, éû˝Dõðí¬˙rÖ

QO, B;>áB°çÞ®2íߤ áá×ç혬i, áá5,uÊB`X•~í

v`´-B, 1/´d_Gøš÷=Bí“, 5´Bø–°z, Bn?DÊ×ù5(A

ôˇì'ª, ß‚ËR¬,û˝F ááôÙ Ï| ØÌ Ùëj "Ô- ÙC]

v

ðå Ù2 ápg s?Ê, ×çÞº5bí´-, éB¥û¬í'Ç-, 5bø–

|Yí%ð, uBø2ä|1ßí[

|(, B;>áBÞ·2|½bíðA, ~~ yy, áá5b,uXMB, éB1B;1

íÍ, 6áá5b,u:KíT, éB.Ûb>ç‘í98, ªJ-Ìê˛íêAç“,

?Dç5bíoä, Böí'« áácc __, çBƒð*7ív`, ,uªJ*5

b™,>§ƒðêíÅÊ, 'òEªJø_1Åíðê

¥¹díêA, uBÞ·2½bíøÜ, cø¤d.#%6Œ¬BíAb

vi

Contents

` . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . i

Abstract . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . iii

Acknowledgement . . . . . . . . . . . . . . . . . . . . . . . . . . . . . v

Contents . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . vii

List of Tables . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . ix

List of Figures . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . xi

1 Introduction 1

2 An UWB 3-dB Directional Coupler 5

2.1 Introduction . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 6

2.2 Theory . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 8

2.2.1 Single-Section Directional Coupler . . . . . . . . . . . . 8

2.2.2 Multisection Directional Couplers . . . . . . . . . . . . 15

2.3 Analysis of the Vertically Installed Planar (VIP) structure 17

2.4 Design Procedure and Realization . . . . . . . . . . . . . . . . 19

2.4.1 Section 1 and Section 5 (Coupling= 15.92dB) . . . . . 21

2.4.2 Section 2 and Section 4 (Coupling= 7.54dB) . . . . . . 23

vii

2.4.3 Section 3 (Coupling= 0.82dB) . . . . . . . . . . . . . . . 26

2.4.4 The total cascaded circuit . . . . . . . . . . . . . . . . . 30

2.5 Compensated VIP structure . . . . . . . . . . . . . . . . . . . . 32

2.6 Measurement . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 38

3 UWB Subharmonic Mixers 43

3.1 Introduction . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 44

3.2 Theory of a Diode Mixer . . . . . . . . . . . . . . . . . . . . . . 45

3.3 The Proposed UWB Subharmonic Quadrature-IF Mixer . . 47

3.3.1 Introduction . . . . . . . . . . . . . . . . . . . . . . . . . . 47

3.3.2 Circuit Realization and Measurements . . . . . . . . . 49

3.4 The Proposed UWB Subharmonic Image Rejection Mixer . 52

3.4.1 Introduction . . . . . . . . . . . . . . . . . . . . . . . . . . 52

3.4.2 Circuit Realization and Measurements . . . . . . . . . 56

4 Conclusion 63

viii

List of Tables

2.1 Table of parameters of symmetrical TEM coupled transmission line

directional couplers . . . . . . . . . . . . . . . . . . . . . . . . . . . 20

2.2 Table of parameters of five-section symmetrical TEM directional cou-

plers . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 21

2.3 Physical dimensions of proposed five -section compensated direc-

tional coupler . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 40

ix

x

List of Figures

2.1 A single-section directional coupler . . . . . . . . . . . . . . . . . . 9

2.2 A directional coupler excited by even-mode sources . . . . . . . . . 9

2.3 A directional coupler excited by odd-mode sources . . . . . . . . . . 10

2.4 The simplified even-mode equivalent circuit . . . . . . . . . . . . . . 10

2.5 The simplified odd-mode equivalent circuit . . . . . . . . . . . . . . 11

2.6 Typical variation of coupling in a single section TEM coupler . . . . 14

2.7 An N-section asymmetrical parallel-coupled multisection directional

coupler . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 15

2.8 An N-section symmetrical parallel-coupled multisection directional

coupler . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 16

2.9 Structure of a VIP directional coupler . . . . . . . . . . . . . . . . . 18

2.10 (a)Even- and (b)odd-mode equivalent circuits of VIP structure . . . 18

2.11 A five-section symmetrical coupler . . . . . . . . . . . . . . . . . . . 22

2.12 (a)An ideal five-section symmetrical coupler and (b)its ideal response 22

2.13 A conventional parallel coupled line coupler . . . . . . . . . . . . . 23

2.14 Simulated results of section 1 and section 5 . . . . . . . . . . . . . . 24

2.15 Cross-sectional view of the type-I VIP coupler . . . . . . . . . . . . 24

xi

2.16 (a)Even- and (b)odd-mode equivalent circuits of the type-I VIP coupler 25

2.17 Even- and odd-mode characteristic impedances versus VIP metal

height (Hmetal) of the type-I VIP coupler with G = 28 mils . . . . . 26

2.18 Simulated results of section 2 and section 4 . . . . . . . . . . . . . . 27

2.19 Cross-sectional view of the type-II VIP coupler . . . . . . . . . . . . 28

2.20 (a)Even- and (b)odd-mode equivalent circuits of the type-II VIP cou-

pler . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 28

2.21 Even- and odd-mode characteristic impedances versus Wg and Hmetal

of the type-II VIP coupler . . . . . . . . . . . . . . . . . . . . . . . 29

2.22 Simulated results of section 3 . . . . . . . . . . . . . . . . . . . . . 30

2.23 Total cascaded circuit in Microwave Office . . . . . . . . . . . . . . 31

2.24 Simulated results of total cascaded circuit in Microwave Office . . . 31

2.25 3-D structure of the five-section directional coupler . . . . . . . . . 32

2.26 Simulated results of total cascaded circuit in HFSS . . . . . . . . . 33

2.27 A compensated VIP coupler . . . . . . . . . . . . . . . . . . . . . . 34

2.28 Top view of compensated VIP coupler . . . . . . . . . . . . . . . . 35

2.29 (a) Top and (b) cross-sectional view of modified compensated VIP

coupler . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 35

2.30 Coupling (S31) of compensated structure with (a) L = 20mil and (b)

W = 20mil . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 36

2.31 Return loss (S11) of compensated structure with (a)L = 20mil and

(b) W = 20mil . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 37

xii

2.32 3-D structure of the five-section compensated directional coupler . . 38

2.33 Simulated results of the five-section compensated directional coupler

in HFSS . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 39

2.34 Photograph of the fabricated five-section 3-dB directional coupler . 39

2.35 Measured responses of the proposed hybrid . . . . . . . . . . . . . . 40

2.36 Compare between measured and simulated responses . . . . . . . . 41

2.37 Measured and simulate amplitude errors of the proposed directional

coupler . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 41

2.38 Measured and simulate phase errors of the proposed directional coupler 42

3.1 Topology of subharmonic IRM . . . . . . . . . . . . . . . . . . . . . 45

3.2 (a) I-V curve of a Schottky diode (b) transconductance waveform of

Schottky diode . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 46

3.3 (a)Topology of UWB subharmonic quadrature-IF mixer (b)Diode

current configuration . . . . . . . . . . . . . . . . . . . . . . . . . . 48

3.4 Photograph of the proposed UWB subharmonic quadrature-IF mixer 50

3.5 UWB subharmonic quadrature-IF mixer circuit configuration . . . . 50

3.6 Time domain wave form of quadrature-IF signal . . . . . . . . . . . 51

3.7 Conversion loss of quadrature-IF mixer . . . . . . . . . . . . . . . . 52

3.8 I/Q amplitude deviation of quadrature-IF mixer . . . . . . . . . . . 53

3.9 Quadrature phase deviation of quadrature-IF mixer . . . . . . . . . 53

3.10 (a) Topology of UWB subharmonic IRM (b) Diode current configu-

ration . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 54

xiii

3.11 Photograph of the proposed UWB subharmonic IRM . . . . . . . . 57

3.12 UWB subharmonic IRM circuit configuration . . . . . . . . . . . . 58

3.13 Conversion loss versus local power of IRM . . . . . . . . . . . . . . 58

3.14 Conversion loss versus RF frequency of IRM . . . . . . . . . . . . . 59

3.15 Image rejection ratio versus RF frequency of IRM . . . . . . . . . . 60

3.16 Conversion loss versus IF frequency of IRM . . . . . . . . . . . . . 60

3.17 Image rejection ratio versus IF frequency of IRM . . . . . . . . . . 61

3.18 Isolation of LO to RF versus RF frequency of IRM . . . . . . . . . 61

xiv

Chapter 1

Introduction

1

Ultra-wideband (UWB) has received a lot of attentions in the wireless communi-

cation applications. It was first conceived in the 1960s and used for radar, sensing,

and military communications in the past 20 years. The FCC opened 3.1-10.6 GHz

available for UWB applications in 2002. UWB systems are focused on providing a

low power, low cost, and wideband performance in a short distance. UWB systems

can often encompass multiple gigaHertz of bandwidth, which poses some interesting

problems to the system engineer. One of the most important is the need to reject

the image frequency to avoid corruption of data or noise problems. The techniques

used to realize different circuit components in a UWB transceiver are quite different

from those proposed in narrow bandwidth radio frequency technology.

One of the key elements in a UWB transceiver is the up- and down-conversion

mixer. It is of interest to find a suitable mixer topology that can achieve good

wideband performance in UWB systems. A wideband, large power, and stable local

oscillator is expensive and hard to obtain. Thus, utilizing a subharmonic mixer to

mix RF and LO signal becomes a better choice due to its halved bandwidth.

There are two general techniques for image rejection. The first one is a pres-

elected band pass filter, which is used to select the desired RF signal and reject

the image one. The amount of image attenuation is strongly depends on the BPF

design. If the IF frequency is quite small compared to LO and RF frequency, design

a suitable BPF will be a hard task to the system designer. For broadband system

such as UWB system, it is very difficult to built an IRM using filter.

The second image rejection technique is known as phase-type. In this method

2

two mixing path with quadrature phase is provided and the properties of in phase

and quadrature paths is exploited to attenuate the image. There is another ben-

efit in this configuration such as improving the linearity by increasing the IP3 in

comparison to a single mixer.

A UWB subharmonic image rejection mixer (IRM) is proposed in this thesis,

which the operation frequency is about 3-13.5GHz, totally covered the UWB band-

width. The most common topology of subharmonic mixer was reported in [1] [2].

Although good performances were achieved, the topology mentioned above all need

a Wilkinson power divider, two Anti Parallel Diode Pairs (APDP), a RF directional

coupler, and an IF directional coupler. The subharmonic mixer proposed here only

needs two diodes, a RF and IF directional coupler. However, the RF/LO quadrature

hybrid has to be ultra-wideband.

In this thesis, first, we will introduce the design procedure of the ultra-broadband

RF/LO quadrature hybrid. Then improvement methods for return loss and isolation

will be discussed. Finally, the proposed topology of UWB subharmonic mixer will

be reported.

3

4

Chapter 2

An UWB 3-dB DirectionalCoupler

5

2.1 Introduction

A 3-dB quadrature coupler is an important and fundamental component in com-

munication system. Nowadays, 3-dB quadrature couplers are widely used in many

microwave circuits, such as low noise amplifiers, phase shifters, balanced mixers.

Branch line coupler is an easy way of realizing the 3-dB quadrature coupler, but

this method requires more space on the print circuit board and is very narrow-

band. A conventional coupler which consists of two parallel microstrip coupled

lines is another method of 3-dB quadrature coupler realization. This method solve

the problem of large area. However, it still has the trouble with the narrow gap

between the two strips, due to its inherent weak coupling nature.

By properly choosing the even- and odd-mode impedances of the two identical

microstrip coupled lines as shown in Fig. 2.1, a four-port directional coupler can

be obtained. Because of the coupling taking place in the backward direction, the

coupler is also called a backward-wave directional coupler. The coupling value varies

with frequency because the electrical length of the coupled line varies with respect

to frequency. Thus, the varying coupling value causes some serious problems when

we want to use a single section coupler in wide-band applications.

A wider bandwidth can be achieved by multisection directional couplers. There

was a complete analysis [3] for multisection directional couplers and the design

had tabled into even- and odd-mode impedance for each section. There are other

papers [4] [5] mentioned that the ultra-wideband performance can also be achieved

by using multisection nonuniform cascaded couplers, but the length of nonuniform

6

multisection coupler is much longer than the uniform one. Besides, nonuniform

couplers have some problems in complicated analysis. The extremely tight coupling

at the center portion of the coupler also limits the practical implementation.

Either uniform or nonuniform couplers have the problem with tight coupling,

several methods have been proposed to solve the coupling problem in the litera-

tures. One method is to use interdigital layout to achieve the tight coupling [6] [7].

The coupling value may higher than conventional coupled line, but still difficult

to implement a coupler tighter than -3dB. Another approach is re-entrant struc-

ture [8] [9] [10], it can achieve tight coupling but hard to fabricate. Tandem cou-

plers [11] might realize the extremely tight coupling section for the multisection

directional couplers. Although this method is easy to implement, but the tandem

couplers take large circuit area and have strong junction discontinuity effect.

Vertically installed planar (VIP) structure is a method of solving the tight cou-

pling problem. It was first proposed in [12] [13]. The VIP structure is composed

of two substrates, the horizontal main substrate and the vertical (VIP) substrate

which vertically installed on the horizontal one. The structure takes the advantages

of easy fabrication, minimum layout area and low cost.

This chapter is organized as follow. First the theory of coupler is discussed.

Then multisection coupler and VIP structure are introduced. The design procedure

of UWB multisection directional coupler, which utilizes VIP structure, is presented

next. Finally, a compensating structure of VIP multisection directional coupler for

return loss improvement is reported.

7

2.2 Theory

Because we want to fabricate a ultra-broadband directional coupler suitable for

UWB subharmonic mixer, the coupler must be operated over broader than 3-10GHz.

According to [3], we must have a five-section TEM mode couplers, then the passband

±0.45dB ripple level and frequency bandwidth 1.5-13.5GHz can be realized. The

1.5 GHz low side frequency is due to the requirement of subharmonic LO.

2.2.1 Single-Section Directional Coupler

A directional coupler has four ports, which are labeled as ”input,” ”direct,” ”cou-

pled,” and ”isolated.” Three important parameters that characterize a directional

coupler are its coupling and directivity, defined as below:

Coupling(dB) = 10 logP1

P3

(2.1)

Directivity(dB) = 10 logP3

P4

(2.2)

Isolation(dB) = 10 logP1

P4

(2.3)

where P1 is the power input at port 1, P3 and P4 are the power output at port 3 and

port 4, respectively. In an ideal case, there is no power transmitted out at isolation

port; but in practice, a small amount of power is always coupled to this port.

The schematic of a four port directional coupler is shown in Fig. 2.1. Since the

coupler is a symmetrical structure, we can analyze the circuit by utilizing even- and

odd-mode excitation. Fig. 2.2 and Fig. 2.3 are figures of the even- and odd-mode

excitation, respectively.

8

Z0

Z0 Z0

Z0

Z0o, Z0e

1

43

2Input port

Direct port

Isolated port

Coupled port

+1V-

Z0

Z0 Z0

Z0

Z0o

1

+1V-

Electric wall (short circuit)

43

2

+1V-

Z0

Z0 Z0

Z0

Z0e3

-1V+

Magnetic wall (open circuit)

4

21

Figure 2.1: A single-section directional coupler

Z0

Z0 Z0

Z0

Z0o, Z0e

1

43

2Input port

Direct port

Isolated port

Coupled port

+1V-

Z0

Z0 Z0

Z0

Z0o

1

+1V-

Electric wall (short circuit)

43

2

+1V-

Z0

Z0 Z0

Z0

Z0e3

-1V+

Magnetic wall (open circuit)

4

21

Figure 2.2: A directional coupler excited by even-mode sources

9

Z0

Z0 Z0

Z0

Z0o, Z0e

1

43

2Input port

Direct port

Isolated port

Coupled port

+1V-

Z0

Z0 Z0

Z0

Z0o

1

+1V-

Electric wall (short circuit)

43

2

+1V-

Z0

Z0 Z0

Z0

Z0e3

-1V+

Magnetic wall (open circuit)

4

21

Figure 2.3: A directional coupler excited by odd-mode sources

+2V-

Z0

Z0 Z0

Z0θo, θe

Z0o, Z0e

1

43

2

+1V-

Z0

Z0 Z0

Z0

Z0o

1

43

2

+1V-

Electric wall (short circuit)

+1V-

Z0

Z0 Z0

Z0

Z0e

1

43

2

-1V+

Magnetic wall (open circuit)

Z0 Z0

θo

1 2Z0o Z0 Z0

θe

1 2Z0e

Figure 2.4: The simplified even-mode equivalent circuit

Because the single section directional coupler is a symmetry structure, we can

simplify the four-port network to a two-port network, as shown in Fig. 2.4 and Fig.

2.5. The ABCD matrices for the even- and odd-mode equivalent circuit are given

by Ae Be

Ce De

=

cos θe jZ0e sin θe

jY0e sin θe cos θe

(2.4a)

and Ao Bo

Co Do

=

cos θo jZ0o sin θo

jY0o sin θo cos θo

(2.4b)

10

+2V-

Z0

Z0 Z0

Z0θo, θe

Z0o, Z0e

1

43

2

+1V-

Z0

Z0 Z0

Z0

Z0o

1

43

2

+1V-

Electric wall (short circuit)

+1V-

Z0

Z0 Z0

Z0

Z0e

1

43

2

-1V+

Magnetic wall (open circuit)

Z0 Z0

θo

1 2Z0o Z0 Z0

θe

1 2Z0e

Figure 2.5: The simplified odd-mode equivalent circuit

The following relationship is useful when conversing the ABCD parameters to

scattering parameters

S11 =A + B/Zo − CZo −D

A + B/Zo + CZo + D(2.5a)

S12 =2(AD −BC)

A + B/Zo + CZo + D(2.5b)

S21 =2

A + B/Zo + CZo + D(2.5c)

S22 =−A + B/Zo − CZo + D

A + B/Zo + CZo + D(2.5d)

thus, the scattering parameters of simplified even- and odd-mode equivalent circuits

can be obtained. According to (2.5), and

S11 =S11e + S11o

2(2.6a)

S21 =S21e + S21o

2(2.6b)

S31 =S21e − S11o

2(2.6c)

S41 =S21e − S21o

2(2.6d)

we can obtain the scattering parameters of the single section directional coupler.

11

After computiong the scattering matrix, the return loss (S11) will be

S11 =(Z0e

2Z0o2 − Zo

4) sin θe sin θo[(Z0

2 + Z0e2) sin θe − 2jZ0Z0e cos θe

] [(Z0

2 + Z0o2) sin θo − 2jZ0Z0o cos θo

]+

j(Z03Z0e − Z0Z0eZ0o

2) cos θe sin θo + j(Z03Z0o − Z0Z0oZ0e

2) sin θe cos θo[(Z0

2 + Z0e2) sin θe − 2jZ0Z0e cos θe

] [(Z0

2 + Z0o2) sin θo − 2jZ0Z0o cos θo

](2.7)

For an ideal parallel TEM coupler, the return loss should be zero. We can see

if θe = θo and

Z0eZ0o = Z02 (2.8)

then (2.7) will identical to zero. (2.8) is a well-known condition of ideal parallel

TEM coupler. If (2.8) is satisfied, the scattering parameters of the four pout network

are given by

S11 = S22 = S33 = S44 = 0 (2.9)

S14 = S41 = S23 = S32 = 0 (2.10)

S12 = S21 = S34 = S43 = S21e =

√1− k2

√1− k2 cos θ + j sin θ

(2.11)

S13 = S31 = S24 = S42 = S11e =jk sin θ√

1− k2 cos θ + j sin θ(2.12)

where S11e and S12e denote the reflection and transmission coefficients of the coupled

lines for the case of even-mode excitation, and θ = βl denotes the electrical length

of the coupler. Coupling coefficient k is given by

k =Z0e − Z0o

Z0e + Z0o

(2.13)

for the coupling is given as C dB (where C is positive quantity), then k is related

to C as

k = 10−C/20 (2.14)

12

The amount of coupling between ports 1 and 3 varies with the electrical length

θ, and the maximum occurs when

θ = βl =π

2rads (2.15)

or

l =π

2β=

λg

4(2.16)

Besides, frequency bandwidth B and frequency bandwidth ratio w are also im-

portant parameters of a directional coupler, which are defined as

B =f2

f1

(2.17)

w =f2 − f1

f0

(2.18)

where

f0 =f2 + f1

2(2.19)

is the center frequency of the coupler and f2 and f1 are the upper and lower fre-

quencies when the coupling equals C + δ, as shown in Fig. 2.4. δ is the tolerance

amount and can be arbitrarily specified.

The frequency bandwidth ratio B and frequency bandwidth ratio w are related

by

w = 2B − 1

B + 1(2.20)

and

B =1 +

w

2

1− w

2

(2.21)

13

Z0

Z0 Z0

Z0

Z0o, Z0e

1

43

2Input port

Direct port

Isolated port

Coupled port

f0 f2f1

C+δ

C-δC

Cou

plin

g (d

B)

Frequency

Figure 2.6: Typical variation of coupling in a single section TEM coupler

From (2.13), we can write Z0e/Z0o in terms of coupling coefficient k as

Z0e

Z0o

=1 + k

1− k(2.22)

further, combine this equation with (2.8), we can get

Z0e = Z0

√1 + k

1− k(2.23)

and

Z0o = Z0

√1− k

1 + k(2.24)

When we want to design a coupler with a given coupling coefficient k and char-

acteristic impedance Z0, we first determine the even- and odd-mode impedances

using (2.23) and (2.24), respectively. The physical length l of the coupler is chosen

14

Port 1 Section1 Section2 Section3 Section2 Section1

Port 4Port 2

Port 3

Z0

Z0

Z0

Z0Z0o1

1

4

3

2Input port

Coupled port

Direct port

Isolated port

………

………

Z0e1

Z0o2Z0e2

Z0o3Z0e3

Z0oNZ0eN

i = 1 i = 2 i = 3i = N

Z0

Z0 Z0

Z0

Z0o1

1

43

2

Input port

Coupled port

Direct port

Isolated port

………

………

Z0e1

Z0oNZ0eN

i = 1 i = (N+1)/2 i = N

………

………

Z0 Z0

………

i = 1 i = (N+1)/2 i = N

………

1 2Z0e1 Z0eN

S11e

Figure 2.7: An N-section asymmetrical parallel-coupled multisection directionalcoupler

as

l =λg

4(2.25)

where λg is the wavelength of the TEM wave in the transmission line medium at

the design frequency f0.

2.2.2 Multisection Directional Couplers

A single-section directional coupler has the benefit of easy fabrication, but it is

hard to obtain an ultra-wideband performance. In order to achieve a near-constant

coupling over a wider frequency bandwidth, a number of coupled sections must be

cascaded [3]. Each section is quarter-wave long at the center frequency. Fig. 2.7

shows an N-section asymmetrical parallel-coupled directional coupler. A multisec-

tion coupler can be either symmetrical or asymmetrical. The term symmetrical

means that a coupler has end-to-end symmetry and employs an odd number of

15

Port 1 Section1 Section2 Section3 Section2 Section1

Port 4Port 2

Port 3

Z0

Z0

Z0

Z0Z0o1

1

4

3

2Input port

Coupled port

Direct port

Isolated port

………

………

Z0e1

Z0o2Z0e2

Z0o3Z0e3

Z0oNZ0eN

i = 1 i = 2 i = 3i = N

Z0

Z0 Z0

Z0

Z0o1

1

43

2

Input port

Coupled port

Direct port

Isolated port

………

………

Z0e1

Z0oNZ0eN

i = 1 i = (N+1)/2 i = N

………

………

Z0 Z0

………

i = 1 i = (N+1)/2 i = N

………

1 2Z0e1 Z0eN

S11e

Figure 2.8: An N-section symmetrical parallel-coupled multisection directional cou-pler

sections. The ith section will be identical to the N + 1− ith section in an N-section

symmetrical coupler as shown in Fig. 2.8. The coupler is referred to as an asym-

metrical directional coupler if it does not have the end-to-end symmetry (Fig. 2.7).

The main difference between symmetrical and asymmetrical coupler is the phase

property. For a symmetrical directional coupler, the signal coupled to the coupled

port has 90 degree phase difference with the signal coupled to the direct port, i.e.

∠S31 = ∠S21 + 90o. The phase relationship has a significant property that it is

independent of frequency. Due to this property, applications for which symmetrical

couplers are best suited including multiplexers, directional filters, balanced mixers,

phase shifters, diplexers, and others to which the 90 degree difference property is

essential. A additional advantage of symmetrical couplers is that the strongest cou-

pling section is in the center and not at one end, so that it becomes less difficult

to connect to all four ports. The asymmetrical couplers do not show the phase

property of symmetrical couplers and are generally used where the couplers are

designed to obtain broadband power division only. Other typical advantages may

16

include higher isolation and better input VSWR.

One of the major limitations of a multisection coupler is that the coupling of the

center section is much tighter than the overall coupling. This can create fabrication

problems in microstrip technology where it is difficult to achieve tight coupling.

Further, different phase velocities of even- and odd-mode equivalent circuits will

degrade the coupler performance. We will use the VIP structure to solve the men-

tioned problems.

2.3 Analysis of the Vertically Installed Planar (VIP)

structure

A VIP circuit consists of broadside coupled lines vertically installed on a main

substrate. A typical directional coupler using a VIP structure is shown in Fig. 2.9.

The VIP substrate is installed on the main substrates. The metal patterns with

coupled lines are made on both sides of the VIP substrate. The gray portions in

Fig. 2.10 are transmission lines. Through conventional parallel coupled line, it is

not easy to have strong coupling between two adjacent microstrip lines. On the

other hand, in the VIP configuration, the vertical substrate contributes to a strong

coupling.

We can assume that the propagation is in the quasi-TEM mode, so that the

capacitance between signal line and ground is the most important quantity. The

effective dielectric constant and the characteristic impedance are computed as

εe =C

C0

(2.26)

17

1

3

4

2

(a) VIP structure (b) Top view (c) Side view

1

2

3

4

Figure 2.9: Structure of a VIP directional coupler

1

3

4

2

(a) VIP structure (b) Top view (c) Side view

1

2

3

4

Magnetic wall Electric wall

(a) (b)

Ca Ca

2Cb

Figure 2.10: (a)Even- and (b)odd-mode equivalent circuits of VIP structure

Z0 =1

√εecC0

(2.27)

where c is the light velocity in the free space, and C and C0 are the capacitance

between the signal line and ground with and without dielectric substrate, respec-

tively.

Now we analyze the VIP structure with the even- and odd-mode equivalent

circuits, as shown in Fig. 2.10. The magnetic wall (even-mode) and electric wall

(odd-mode) are located at the symmetry planes of the VIP substrate, respectively.

18

The magnetic wall acts like an open circuit and the electric wall acts like an short

circuit. The even-mode equivalent capacitor contains only Ca, but the odd-mode

equivalent capacitor contains Ca and 2Cb. Because the distance (4mil) between the

symmetry plane and the conductor is very small, 2Cb becomes the major term by

its large value. The odd-mode equivalent capacitance, which contains 2Cb, is much

larger than the even-mode equivalent capacitance. Thus, by (2.26) and (2.27), the

odd-mode characteristic impedance (Z0o) will be much smaller than the even-mode

characteristic impedance (Z0e). We can adjust the heights of the metal patterns on

the VIP substrate to fit the desired even- and odd-mode characteristic impedance.

In the odd-mode excitation, due to the major capacitance 2Cb, the electric filed

is mostly confined in the VIP substrate (εr = 3.38). In the even-mode excitation,

most of the electric field is in the air (εr = 1). Therefore, the VIP structure has

inherently different even- and odd-mode phase velocities. The phase velocity of odd-

mode is slower than the even-mode, this will cause poor isolation and directivity.

We will put the dielectric blocks, which are the same materials as the main and

VIP substrate, to solve the problem. After adding the dielectric blocks, the phase

velocity of even- and odd-mode can be nearly equal.

2.4 Design Procedure and Realization

We want to design a symmetrical multisection TEM coupler with the following

specifications:

Center frequency = 7.5GHz

19

δ Z1 Z2 Z3 w B0.05 1.05792 1.32624 3.81243 1.20488 4.030710.10 1.07851 1.37268 3.97615 1.32559 4.931140.15 1.09451 1.40890 4.10191 1.39889 5.654370.20 1.10921 1.44029 4.21023 1.45184 6.297140.25 1.12314 1.46883 4.30864 1.49333 6.894740.30 1.13659 1.49551 4.40089 1.52744 7.464620.35 1.13973 1.52091 4.48917 1.55639 8.016980.40 1.16266 1.54541 4.57491 1.58152 8.558450.45 1.17547 1.56926 4.65912 1.60371 9.093670.50 1.18822 1.59265 4.74253 1.62357 9.626090.60 1.21370 1.63864 4.90924 1.65791 10.692920.70 1.23941 1.68425 5.07867 1.68691 11.775680.80 1.26555 1.73013 5.25363 1.71196 12.887200.90 1.29235 1.77678 5.43655 1.73402 14.038601.00 1.31998 1.82466 5.62978 1.75370 15.24047

Table 2.1: Table of parameters of symmetrical TEM coupled transmission linedirectional couplers

Mean coupling = 3dB

Maximal ripple level = ±0.45dB

Operating frequency = 1.5− 13.5GHz

Frequency bandwidth ratio = 9

Cristal and Young had generated design tables for equal-ripple symmetrical

couplers [3], the design table is reproduced in Table 2.1. We use the Table 2.1.

to design a symmetrical coupler. It can be found that a 3-dB, five-section coupler

designed for a ripple level of ±0.45dB exhibits a frequency bandwidth ratio B =

9.09367. Because this figure of B is very close to the specified value of B = 9, it is

sufficient to employ a five-section coupler to achieved the desired specifications.

20

Section 1,5 Section2,4 Section3Z0e(Ω) 58.77 78.46 232.96Z0o(Ω) 42.54 31.86 10.73

k 0.16 0.42 0.91Coupling(dB) 15.92 7.54 0.82

δ(dB) 0.45w 1.60B 9.1

Table 2.2: Table of parameters of five-section symmetrical TEM directional couplers

The parameters in Table 2.1 are the normalized values of even-mode charac-

teristic impedance. We can multiply the parameters by 50 to obtain the actual

even-mode characteristic impedance. After knowing the Z0e of each section, Z0o,

k and coupling in dB can be found by (2.8), (2.13) and (2.14). The values of Z0e,

Z0o, and coupling for the desired five-section symmetrical coupler are given in Ta-

ble 2.2. The figure of five-section symmetrical coupler is shown in Fig. 2.11. Fig.

2.12 is an ideal five-section symetrical coupler and its ideal response, which is sim-

ulated by Microwave Office. Note that each section is quarter-wave long at the

center frequency. Section 1 and section 5 can be fabricated by parallel coupled line

coupler. But if section 2 and section 3 are implemented by the same method, the

gap between the coupled lines will be 1.78mil and 0.066mil, respectively, which are

impractical designs. VIP structure may be a better choice to implement the tight

coupling section.

2.4.1 Section 1 and Section 5 (Coupling= 15.92dB)

From Table 2.2, the section 1 and section 5 are loosly coupled sections. The con-

ventional parallel coupled line coupler is suitable for realizing these two sections.

21

Section1 Section2 Section3 Section4 Section5

Z0

Z0

Z0

Z0Z0o1

1

4

3

2Input port

Coupled port

Direct port

Isolated port

………

………

Z0e1

Z0o2Z0e2

Z0o3Z0e3

Z0oNZ0eN

i = 1 i = 2 i = 3i = N

Z0

Z0 Z0

Z0

Z0o1

1

43

2

Input port

Coupled port

Direct port

Isolated port

………

………

Z0e1

Z0oNZ0eN

i = 1 i = (N+1)/2 i = N

………

………

Z0 Z0

………

i = 1 i = (N+1)/2 i = N

………

1 2Z0e1 Z0eN

S11e

Input port

Direct port

Coupled port

Isolated port

Figure 2.11: A five-section symmetrical coupler

(dB

)

S11 S41S31S21

0

-20

-40

-60

-80

-100

-120

Frequency (GHz)

15107.52.5 12.55

(a)

(b)

Figure 2.12: (a)An ideal five-section symmetrical coupler and (b)its ideal response

22

1

3

4

2

(a) VIP structure (b) Top view (c) Side view

1

2

3

4

Magnetic wall Electric wall

(a) (b)

Ca Ca

2Cb

width

gaplength

Figure 2.13: A conventional parallel coupled line coupler

The parallel coupled line coupler is shown in Fig. 2.13.

The specifications of section 1 and section 5 are Z0e = 58.77Ω, Z0o = 42.54Ω,and

Coupling = 15.92dB. Roughly calculating the gap, width and the length of coupled

line, we obtained the gap = 15mil, width = 40mil, length = 233mil. By using

the EM simulator such as Ansoft HFSS to simulate circuit, the simulated results

of scattering parameters is shown in Fig. 2.14. The Coupling (S31) in Fig. 2.14

is 15.53dB, which is very close to the specified value, and the center frequency is

indeed at 7.5GHz.

2.4.2 Section 2 and Section 4 (Coupling= 7.54dB)

The Coupling of section 2 and section 4 should be 7.54dB, this cannot be imple-

mented by the conventional parallel coupled line coupler due to physical limitation.

There are two different types of VIP coupler were proposed in [14]. A type-I

VIP coupler is shown in Fig. 2.15. There are four metal strips of which two are

on the main substrate and the other two are on the VIP substrate. The strips

23

10

0

-10

-20

-30

-40

-50

(dB

)

Frequency (GHz)

15107.52.5 12.55

S11 S41S31S21

10

0

-10

-20

-30

-40

-50

(dB

)

Frequency (GHz)

15107.52.5 12.55

S11 S41S31S21

0

-5

-10

-15

-20

-25

-30

(dB

)

S11 S41S31S21

Frequency (GHz)

15107.52.5 12.55

S21

Figure 2.14: Simulated results of section 1 and section 5

Metal

G

Hm

etal

H VIP

Dielectric substrate

W

Metal

WG

Hm

etal

H VIP

Dielectric substrate

Wgnd

WVIP

WVIP

Magnetic wall Electric wall

(a) (b)

Ca Ca

CcCc

Ca

Electric wall

(a) (b)

Ca

2Cb

Ca

Magnetic wall

CaCa

2Cb

2Cb

Figure 2.15: Cross-sectional view of the type-I VIP coupler

24

Metal

G

Hm

etal

H VIP

Dielectric substrate

W

Metal

WG

Hm

etal

H VIP

Dielectric substrate

Wgnd

WVIP

WVIP

Magnetic wall Electric wall

(a) (b)

Ca Ca

CcCc

Ca

Electric wall

(a) (b)

Ca

2Cb

Ca

Magnetic wall

CaCa

2Cb

2Cb

Figure 2.16: (a)Even- and (b)odd-mode equivalent circuits of the type-I VIP coupler

on the main and VIP substrates are connected at two ends of the coupler. The

performance of the coupler is improved by adding dielectric blocks at both sides

of the VIP substrate, which use the same material as the main substrate and VIP

substrate. This coupler can implement a coupler with coupling from moderate to

tight coupling.

The even- and odd-mode equivalent circuits of the type-I VIP coupler is shown

in Fig. 2.16. It shows that the total equivalent capacitance of each equivalent circuit

is the combination of every capacitance shown in the figure. This means that three

degrees of freedom, which are the VIP metal height, the width of the strips on the

main substrate, and the gap width on the main substrate, are available to choose

the even- and odd-mode characteristic impedances. The total width of the coupler

has to be chosen carefully so that the junctions connecting to other sections can be

laid out with minimal discontinuity.

The even- and odd-mode characteristic impedances with respect to the VIP

25

Z0e

Z0o

Z0e

Z0o

Figure 2.17: Even- and odd-mode characteristic impedances versus VIP metalheight (Hmetal) of the type-I VIP coupler with G = 28 mils

metal height and the width of the strips on the main substrate are extracted by the

EM simulator Ansoft HFSS. Fig. 2.17 depicts the extracted data, in which the gap

width between two strips on the main substrate 28 mils.

As given in Fig. 2.17, Hmetal = 7mil, W = 15mil, G = 28mil are chosen for

the case if Z0e = 78.46Ω and Z0o = 31.86Ω. The simulated results of scattering

parameters is shown in Fig. 2.18. The Coupling (S31) is 7.7dB at 7.5GHz, which is

very close to the ideal value 7.54dB.

2.4.3 Section 3 (Coupling= 0.82dB)

The type-I VIP coupler can not implement the extremely tight-coupled center sec-

tion even when W equals 0. To achieve a coupling value as tight as 0.8 dB, the

type-II VIP coupler is proposed in [14], as shown in Fig. 2.19. The ground plane

26

10

0

-10

-20

-30

-40

-50

(dB

)

Frequency (GHz)

15107.52.5 12.55

S31S21

10

0

-10

-20

-30

-40

-50

(dB

)

Frequency (GHz)

15107.52.5 12.55

S11 S41S31S21

0

-5

-10

-15

-20

-25

-30

(dB

)

S11 S41S31S21

Frequency (GHz)

15107.52.5 12.55

S21

S11 S41S31S21

Figure 2.18: Simulated results of section 2 and section 4

of type-II VIP coupler in the main substrate changes to two metal strips. Utilizing

this finite-extent ground plane, the VIP coupler can achieve a coupling tighter than

0.8 dB. Again, a dielectric block as the type-I is used to compensate the modal

phase velocities.

The type-II VIP coupler also has three degrees of freedom to choose the even-

and odd-mode characteristic impedance such as the type-I VIP coupler. The even-

and odd-mode equivalent circuits of the type-II VIP coupler are shown in Fig. 2.20.

The larger the gap between the two strips on the ground plane, the smaller the

equivalent capacitance. Thus, the characteristic impedance of even-mode can be

controlled by the Wg. In addition, the width of ground strips also has influence on

Z0e. The metal height on the VIP substrate can affect the odd-mode characteristic

27

Metal

G

Hm

etal

H VIP

Dielectric substrate

W

Metal

WG

Hm

etal

H VIP

Dielectric substrate

Wgnd

WVIP

WVIP

Magnetic wall Electric wall

(a) (b)

Ca Ca

CcCc

Ca

Electric wall

(a) (b)

Ca

2Cb

Ca

Magnetic wall

CaCa

2Cb

2Cb

Figure 2.19: Cross-sectional view of the type-II VIP coupler

Metal

G

Hm

etal

H VIP

Dielectric substrate

W

Metal

WG

Hm

etal

H VIP

Dielectric substrate

Wgnd

WVIP

WVIP

Magnetic wall Electric wall

(a) (b)

Ca Ca

CcCc

Ca

Electric wall

(a) (b)

Ca

2Cb

Ca

Magnetic wall

CaCa

2Cb

2Cb

Figure 2.20: (a)Even- and (b)odd-mode equivalent circuits of the type-II VIP cou-pler

28

Z0e

Z0o

Z0e

Z0o

Z0e

Z0o

Figure 2.21: Even- and odd-mode characteristic impedances versus Wg and Hmetal

of the type-II VIP coupler

impedance. A large value of height of the metal on the VIP substrate will make

itself and the electric plane forming a giant parallel plate capacitor. This means

that the odd-mode characteristic impedance will significant decrease if the metal

height becomes larger.

The even- and odd-mode characteristic impedances with respect to the Wg and

Hmetal of the type-II VIP coupler are shown in Fig. 2.21. According to this figure,

the Wg, Hmetal, and Wgnd are chosen to be 208mil, 60mil, and 16mil, respectively.

The simulated result is shown in Fig. 2.22. The Coupling in Fig2.22 only has

0.02dB difference to the ideal value.

29

10

0

-10

-20

-30

-40

-50

(dB

)

Frequency (GHz)

15107.52.5 12.55

S11 S41S31S21

10

0

-10

-20

-30

-40

-50

(dB

)

Frequency (GHz)

15107.52.5 12.55

S11 S41S31S21

0

-5

-10

-15

-20

-25

-30

(dB

)

S11 S41S31S21

Frequency (GHz)

15107.52.5 12.55

S21

Figure 2.22: Simulated results of section 3

2.4.4 The total cascaded circuit

After finishing the design of each section of the UWB multisection directional cou-

pler, the scattering parameters of UWB multisection directional coupler are ob-

tained by cascading each section, utilizing a circuit simulator Microwave Office.

The total cascaded circuit in Microwave Office is shown in Fig. 2.23 and simulated

results are shown in Fig2.24.

The S21 and S31 shown in Fig. 2.24 are very close to the S21 and S31 of an ideal

five-section directional coupler, as shown in Fig. 2.12. But the isolation and return

loss of this cascaded circuit are quite different from the ideal value.

Fig. 2.24 shows that the initial designs of each section are dependable. When

each section cascaded one by one, the junction discontinuity must be taken into

30

Frequency (GHz)

15107.52.5 12.55

(dB

)

-20

-30

-10

0

10

S11 S41S31S21

S11 S41S31S21

(dB

)

-30

-40

-20

-10

0

Frequency (GHz)

15107.52.5 12.55

S21 S31

Figure 2.23: Total cascaded circuit in Microwave Office

Frequency (GHz)

15107.52.5 12.55

(dB

)

-20

-30

-10

0

10

S11 S41S31S21

S11 S41S31S21

(dB

)

-30

-40

-20

-10

0

Frequency (GHz)

15107.52.5 12.55

S21 S31

Figure 2.24: Simulated results of total cascaded circuit in Microwave Office

31

S21

S31

S41

S11(dB

)(d

egre

e)

S21 S31

(a)

(b)

Figure 2.25: 3-D structure of the five-section directional coupler

consideration. In order to get more accurate results, the whole five-section direc-

tional coupler is simulated by an EM simulator Ansoft HFSS. The 3-D structure

of the five-section directional coupler is shown in Fig. 2.25, and simulated results

are shown in Fig. 2.26. Fig. 2.26(a) shows that the useful bandwidth of this mul-

tisection directional coupler is from 1.5 to 13.5GHz. The maximal amplitude error

between the coupled port and direct port is about 1.2dB (ideal value is 0.9dB). The

phase of the coupled and direct port has a 90 degree difference due to its symmetric

nature.

Fig. 2.26 shows the performance degradation at high frequency. The main

reason for the degradation is from the junction discontinuity effect between each

section, especially between section 3 and its neighbors. The return loss is -18dB at

10.75GHz, which is not good enough. A compensated VIP structure is useful for

the return loss improvement.

2.5 Compensated VIP structure

In order to improve the performance of a VIP coupler, a new planar compensation

structure has been studied experimentally and theoretically in [15]. By only adding

32

S21

S31

S41

S11(dB

)(d

egre

e)

S21 S31

(a)

(b)

Figure 2.26: Simulated results of total cascaded circuit in HFSS

33

1

2

3

4

W2

L2

L1

W1

L

1 3

42

W

L

Metal

Dielectric substrate

Metal

Dielectric substrate

Preceding section succeeding section

(a)

(b)

Metal

Dielectric substrate

Figure 2.27: A compensated VIP coupler

some simple short microstrip stubs to the VIP coupler circuit, a better frequency

response on the S parameters can be obtained.

A configuration of the proposed compensated VIP directional coupler is illus-

trated in Fig. 2.27. Fig. 2.28 is just a top view of this device. The length L is

chosen as a quarter wavelength at the center frequency. The compensation stubs

(W2 × L2) at the terminals of the coupling sections will influence mainly on the

magnitude of the return loss and isolation, and the compensation stub in the center

will contribute mainly to reduce the difference between the even and odd mode

effective dielectric constants.

A modified topology suitable for the multisection directional coupler is shown

in Fig. 2.29. The center compensation stub is removed since the magnitude of the

isolation and return loss are the major issue.

The results of return loss (S11) and coupling (S31) with respect to the stub width

and length are simulated by Ansoft HFSS, as shown in Fig. 2.30 and 2.31.

34

1

2

3

4

W2

L2

L1

W1

L

1 3

42

W

L

Metal

Dielectric substrate

Metal

Dielectric substrate

Preceding section succeeding section

(a)

(b)

Metal

Dielectric substrate

Figure 2.28: Top view of compensated VIP coupler

1

2

3

4

W2

L2

L1

W1

L

1 3

42

W

L

Metal

Dielectric substrate

Metal

Dielectric substrate

preceding section succeeding section

(a)

(b)

Metal

Dielectric substrate

Figure 2.29: (a) Top and (b) cross-sectional view of modified compensated VIPcoupler

35

(dB

)

-15

-10

0

-5

W=10mil W=20mil W=30mil W=40mil uncompensated

Frequency (GHz)

15107.52.5 12.55

(a)

(dB

)

-15

-10

0

-5

Frequency (GHz)

15107.52.5 12.55

L=10mil L=20mil L=30mil L=40mil uncompensated

(b)

Figure 2.30: Coupling (S31) of compensated structure with (a) L = 20mil and (b)W = 20mil

36

(dB

)

0

-10

-20

-30

-40

-50

-60

Frequency (GHz)

15107.52.5 12.55

W=10mil W=20mil W=30mil W=40mil uncompensated

(a)

Frequency (GHz)

15107.52.5 12.55

(dB

)

0

-10

-20

-30

-40

-50

-60

L=10mil L=20mil L=30mil L=40mil uncompensated

(b)

Figure 2.31: Return loss (S11) of compensated structure with (a)L = 20mil and (b)W = 20mil

37

: compensated stubs

S21S31

S41

S11

Figure 2.32: 3-D structure of the five-section compensated directional coupler

Observing Fig. 2.30, the coupling maintains the same value, no matter how the

width or length changes. Fig. 2.31 shows that the return loss varies with the width

or length. These results show that the terminal stubs only affect the magnitude of

return loss and have no effect on the coupling.

After choosing W = 20mil and L = 20mil, which make return loss having the

lowest value, we adopt the compensated section into the total cascaded circuit to

implement the UWB multisection directional coupler. The 3-D structure of the

compensated multisection coupler is shown in Fig. 2.32. Fig. 2.33 is the simulated

results by utilizing Ansoft HFSS. Compare Fig. 2.33 with the previous result Fig.

2.26, the return loss has a significant improvement by at least 6dB at 7.5-12.5GHz.

The benefits of the compensated structure are the return loss can be effective

suppress and the coupling won’t be affected by adding the compensated stubs.

2.6 Measurement

The photograph of the proposed compensated five-section directional coupler is

shown in Fig. 2.34. The main substrate, VIP substrate, and dielectric blocks are

38

: compensated stubs

S21S31

S41

S11

Figure 2.33: Simulated results of the five-section compensated directional couplerin HFSS

S11 S41S31S21

Frequency (GHz)

15107.52.5 12.55

(dB

)

0

-20

-40

-60

-80

-100

-120

Frequency (GHz)

15107.52.5 12.55

S31 S21 S31 S21

Measurement Simulation

(dB

)

-20

-18

-16

-14

-12-10

-8

-6

-4

-2

0

|S21

|-|S 3

1|(dB

)

-10

-8

-6

-4

-20

2

4

6

8

10

Frequency (GHz)

15107.52.5 12.55

MeasurementSimulation

Phas

e di

ffer

ence

(deg

ree)

-200

-100

0

100

200MeasurementSimulation

Figure 2.34: Photograph of the fabricated five-section 3-dB directional coupler

39

S11 S41S31S21

Frequency (GHz)

15107.52.5 12.55

(dB

)

0

-10

-20

-30

-40

-50

-60

Frequency (GHz)

15107.52.5 12.55

S31 S21 S31 S21

Measurement Simulation

(dB

)

-20

-18

-16

-14

-12-10

-8

-6

-4

-2

0

|S21

|-|S 3

1|(dB

)

-10

-8

-6

-4

-20

2

4

6

8

10

Frequency (GHz)

15107.52.5 12.55

MeasurementSimulation

Phas

e di

ffer

ence

(deg

ree)

-200

-100

0

100

200MeasurementSimulation

Frequency (GHz)

15107.52.5 12.55

Figure 2.35: Measured responses of the proposed hybrid

Section Z0e Z0o W G Hmetal Wg Wgnd L1,5 58.77 42.54 40 15 NA NA NA 2332,4 78.46 31.86 15 28 7 NA NA 2003 232.96 10.73 NA NA 60 208 16 209

units (mil)

Table 2.3: Physical dimensions of proposed five -section compensated directionalcoupler

all Rogers RO4003 with a dielectric constant of 3.58 and thickness of 8, 20, and

60 mils, respectively. Depicted in Figs. 2.35 is the measured result which matches

well with the simulation result as shown in Fig. 2.36. The simulated and measured

amplitude and phase errors are shown in Fig. 2.37 and Fig. 2.38, respectively.

The measured amplitude balances between port 3 (coupled port) and port 2 (direct

port) is less than 2dB, and the phase difference is keeping near 90 degree over the

designed frequency of 1.5-13.5 GHz.

40

S11 S41S31S21

Frequency (GHz)

15107.52.5 12.55

(dB

)

0

-20

-40

-60

-80

-100

-120

Frequency (GHz)

15107.52.5 12.55

S31 S21 S31 S21

Measurement Simulation(d

B)

-20

-18

-16

-14

-12-10

-8

-6

-4

-2

0

|S21

|-|S 3

1|(dB

)

-10

-8

-6

-4

-20

2

4

6

8

10

Frequency (GHz)

15107.52.5 12.55

MeasurementSimulation

Phas

e di

ffer

ence

(deg

ree)

-200

-100

0

100

200MeasurementSimulation

Figure 2.36: Compare between measured and simulated responses

S11 S41S31S21

Frequency (GHz)

15107.52.5 12.55

(dB

)

0

-20

-40

-60

-80

-100

-120

Frequency (GHz)

15107.52.5 12.55

S31 S21 S31 S21

Measurement Simulation

(dB

)

-20

-18

-16

-14

-12-10

-8

-6

-4

-2

0

|S21

|-|S 3

1|(dB

)

-10

-8

-6

-4

-20

2

4

6

8

10

Frequency (GHz)

15107.52.5 12.55

MeasurementSimulation

Phas

e di

ffer

ence

(deg

ree)

-200

-100

0

100

200MeasurementSimulation

Figure 2.37: Measured and simulate amplitude errors of the proposed directionalcoupler

41

S11 S41S31S21

Frequency (GHz)

15107.52.5 12.55

(dB

)

0

-20

-40

-60

-80

-100

-120

Frequency (GHz)

15107.52.5 12.55

S31 S21 S31 S21

Measurement Simulation

(dB

)

-20

-18

-16

-14

-12-10

-8

-6

-4

-2

0

|S21

|-|S 3

1|(dB

)

-10

-8

-6

-4

-20

2

4

6

8

10

Frequency (GHz)

15107.52.5 12.55

MeasurementSimulation

Phas

e di

ffer

ence

(deg

ree)

-200

-100

0

100

200MeasurementSimulation

Frequency (GHz)

15107.52.5 12.55

Figure 2.38: Measured and simulate phase errors of the proposed directional coupler

We restate the design procedure of UWB multisection directional coupler as

follows. According to the design table listed in Table 2.1 or in [3], five-section

coupler are needed to fulfill such specifications. The corresponding even- and odd-

mode characteristic impedances of each section are detailed in Table 2.2. For the

first and fifth sections, use a circuit simulator to get physical dimensions. Then, use

Fig. 2.21 to obtain the physical dimensions of section 3. Last, to design sections

2 and 4, designers should properly choose the width and distance of the strips on

the main substrate by the aid of Fig. 2.17 to minimize the discontinuities between

sections. Finally, the whole directional coupler performances are simulated by 3-D

EM simulator Ansoft HFSS. The physical dimensions of each section are shown in

Table 2.3.

42

Chapter 3

UWB Subharmonic Mixers

43

3.1 Introduction

One of the benefits of designing a system for operation in the milimeter wave band

is the capability to utilize large bandwidths for the transmission of high volumes of

data. How to reject the image frequency to avoid noise and data corruption is the

major issue for an UWB mixer. Whilst in a narrow band system a simple filter can

be chosen to do this, it can be impossible in a wide band system. A way of avoiding

this issue is to incorporate an image reject mixer that, with the use of directional

couplers, can inherently reject the image signal.

Image rejection mixers are becoming more important, since the requirement

for cheaper and smaller are becoming a trend for every 3C products. There are

additional benefits provided by IRM topology to the system designer. First, the

use of two mixers in a balanced configuration increases the IP3 by 3dB from a

single mixer, and thus improving the linearity and allowing higher input RF power

for the same spurious response. This achieves better spurious free dynamic range

in a receiver. Secondly, the system designer can choose to isolate either an upper

or lower sideband simply by modifying which port of IF directional coupler is the

output. Finally, the use of directional couplers to apply the signals to the mixers

inherently gives good port return loss and improved isolations.

The subharmonic mixer can be driven by a local source with half of the LO

frequency. In fact, the RF signal is mixed with the second harmonic of the LO

source. This characteristic is useful in the wideband application since the wideband

local oscillator is hard to obtain. The most common topology of subharmonic image

44

WilkinsonPowerDivider

RF90o

Hybrid

IF90o

Hybrid

LO RF

IF

RF/LO 90º

HybridA

B

LPF

LPF

HP

FH

PF

RF

LO

IF1

IF2

ID1

VRF,A

VRF,B

ID2,RF

ID1,RF

VLO,A

VLO,B

ID2.LO

ID1,LO

ID2

ID2'

(a)

(b)

Subharmonic mixer

Figure 3.1: Topology of subharmonic IRM

rejection mixer was proposed in [1] [2]. The mixer consists of a Wilkinson power

divider, two Anti Parallel Diode Pairs (APDP), a RF directional coupler and an IF

direction coupler, as shown in Fig. 3.1.

The UWB subharmonic quadrature-IF and image rejection mixer proposed in

this thesis can be fabricated without a Wilkinson power divider and only needs two

diodes. It has the benefit of reducing the number of elements.

3.2 Theory of a Diode Mixer

Fig. 3.2(a) shows the I-V curve of a Schottky diode. When applying a local oscillat-

ing (LO) signal on a diode, the signal would be rectified since only the positive cycle

can turn on the diode. Hence, the transconductance waveform gLO(t) of Schottky

diode is shown in Fig. 3.2(b). The Fourier expansion of gLO(t) can be expressed as

gLO(t) =∞∑

n=−∞

gnejnωLOt (3.1)

45

RF/LO 90º

HybridA

B

LPF

LPF

HP

FH

PF

IF 90º

Hybrid

RF

LO

IF1

IF2

ID1

VRF,A

VRF,B

ID2,RF

ID1,RF

VLO,A

VLO,B

ID2.LO

ID1,LO

ID2

ID2'

V

I

t

1/fLO

gLO(t)

VLO(t)

(a)

(b)

(a) (b)

Figure 3.2: (a) I-V curve of a Schottky diode (b) transconductance waveform ofSchottky diode

When RF signal is applied on the diode at the same time, due to the nonlinearity

of the diode, the RF voltage across the diode will contain harmonic terms of the

RF frequency. This RF voltage is shown as

VRF (t) =∞∑

m=−∞

VmejmωRF t (3.2)

thus, the diode current id can be expressed as

id(t) =∞∑

m=−∞

∞∑n=−∞

gnVmej(mωRF +nωLO)t (3.3)

id shown above contains all intermodulation products of the RF frequency and LO

frequency. Fundamental mixing occurs when (m,n) equals (1,-1) or (-1,1). All other

higher order products can be eliminated by a filter.

46

3.3 The Proposed UWB Subharmonic Quadrature-

IF Mixer

3.3.1 Introduction

The proposed topology of UWB subharmonic quadrature-IF mixer is shown in Fig.

3.3(a). The RF/LO 90o hybrid in Fig. 3.3(a) is identical to the UWB multisection

directional coupler proposed in chapter 2. The RF and LO input signal are applied

to the input port and isolated port in Fig. 2.11, respectively. RF signal is isolated

with LO signal due to the property of a directional coupler.

We analyze the proposed mixer by the relationships between diode current and

LO, RF signal. Fig. 3.3(b) shows that the RF signal at point B, which is marked in

Fig. 3.3(a), has a 90 degree phase delay to the RF signal at point A. Here, we only

concerned the phase relationship and ignored the amplitude changing. LO signal

at point A also has a 90 degree phase delay to the LO signal at point B. Thus,

the diode current ID2,RF of RF signal has a 90 degree phase ahead of ID1,RF . The

relation between ID2,RF and ID1,RF can be expressed as

ID2,RF = (j)mID1,RF (3.4)

also, ID1,LO and ID2,LO has the following relationship

ID2,LO = (−j)nID1,LO (3.5)

after combining the equations above, the total relation between ID1 and ID2 can

be shown as

ID2 = (j)m(−j)nID1 (3.6)

47

WilkinsonPowerDivider

RF90o

Hybrid

IF90o

Hybrid

LO RF

IF

RF/LO 90º

HybridA

B

LPF

LPF

HP

FH

PF

RF

LO

IF1

IF2

ID1

VRF,A

VRF,B

ID2,RF

ID1,RF

VLO,A

VLO,B

ID2.LO

ID1,LO

ID2

ID2'

(a)

(b)

Figure 3.3: (a)Topology of UWB subharmonic quadrature-IF mixer (b)Diode cur-rent configuration

48

the index m and n in the equations above represent the harmonics of RF and LO

signal. I ′D2 means the diode current of reversing the direction of ID2

I ′D2 = −ID2 = −(j)m(−j)nID1 (3.7)

IF signal appears when (m,n) equals (1,-2) or (-1,2) in a subharmonic mixer.

For (m,n) equals (1,-2)

I ′D2 = −j(−j)−2ID1 = jID1 (3.8)

where ID1 has a phase delay of 90 degree compared to I ′D2. On the other hand, for

(m,n) equals (-1,2)

I ′D2 = −(j)−1(−j)2ID1 = −jID1 (3.9)

I ′D2 has a phase delay of 90 degree compared to ID1. The above property actually

performed a quadrature-IF mixer.

3.3.2 Circuit Realization and Measurements

Fig. 3.4 and Fig. 3.5 show photograph and configuration of the proposed UWB

mixer. The circuit is fabricated on RO4003 substrate with a dielectric constant of

3.58 and thickness of 20mil. The RF and IF frequency are chosen to be 3-13GHz

and 60MHz, respectively.

The total circuit consists of a UWB RF/LO directional coupler, two diodes, a

low-pass filter, and a high-pass filer. UWB RF/LO directional coupler utilizes the

multisection direction coupler discussed in chapter 2. LPF and HPF are realized

by a simple conductor and capacitor.

49

Figure 3.4: Photograph of the proposed UWB subharmonic quadrature-IF mixer

RF/LO hybrid

Ground pad

LPFHPF

Jump wire

IF hybrid

RF

LOIF

IF

IF

IFLO

RF

Figure 3.5: UWB subharmonic quadrature-IF mixer circuit configuration

50

NCTU-CM MIC LAB

7RFf GHz=

2IF LO RFf f f= −

8RFf GHz=20° 90°

Figure 3.6: Time domain wave form of quadrature-IF signal

Applying sinusoidal waves into RF and LO port, a time-domain voltage wave-

form can be observed by an oscilloscope. Fig. 3.6 is the measured voltage waveform

of proposed mixer under the condition of local power 10dBm, RF frequency 8GHz.

Fig. 3.6 shows that the amplitude deviation is almost zero at this frequency and

phase difference between two output ports is 90 degree.

Fig. 3.7 shows the conversion loss of the quadrature-IF mixer. The conversion

loss is lower than 15dB when local power is larger than 10dBm. The RF and IF

frequency are given at 8GHz and 60MHz, respectively.

51

Local Power (dBm)

0 2 4 6 8 10 12 14 16

Con

vers

ion

Loss

(dB

)

10

15

20

25

30

35

40

RF Frequency (GHz)

2 4 6 8 10 12 14

Isol

atio

n (d

B)

12

14

16

18

20

22

24

26

3.7 3.18

Figure 3.7: Conversion loss of quadrature-IF mixer

Fig. 3.8 and Fig. 3.9 show the I/Q amplitude deviation and quadrature phase

deviation of the quadrature-IF mixer. The amplitude deviation is less than ±3dB

when RF frequency is from 3-13GHz and IF is fixed at 60MHz. The local source

power is set around 10dBm.

3.4 The Proposed UWB Subharmonic Image Re-

jection Mixer

3.4.1 Introduction

The proposed topology of UWB subharmonic image rejection mixer (IRM) is shown

in Fig. 3.10. This image rejection mixer is same as the quadrature-IF mixer dis-

cussed in the previous section except an additional IF hybrid. We utilize the same

method to analyze the proposed mixer.

52

RF Frequency (GHz)

2 4 6 8 10 12 14

Qua

drat

ure

Phas

e D

evia

tion

(dB

)

-80

-60

-40

-20

0

20

40

60

80

RF Frequency (GHz)

2 4 6 8 10 12 14

I/Q A

mpl

itude

Dev

iatio

n (d

B)

-3

-2

-1

0

1

2

3

4

3.9 3.8

Figure 3.8: I/Q amplitude deviation of quadrature-IF mixer

RF Frequency (GHz)

2 4 6 8 10 12 14

Qua

drat

ure

Phas

e D

evia

tion

(dB

)

-80

-60

-40

-20

0

20

40

60

80

RF Frequency (GHz)

2 4 6 8 10 12 14

I/Q A

mpl

itude

Dev

iatio

n (d

B)

-3

-2

-1

0

1

2

3

4

3.9 3.8

Figure 3.9: Quadrature phase deviation of quadrature-IF mixer

53

RF/LO 90º

HybridA

B

LPF

LPF

HP

FH

PF

IF 90º

Hybrid

RF

LO

IF1

IF2

ID1

VRF,A

VRF,B

ID2,RF

ID1,RF

VLO,A

VLO,B

ID2.LO

ID1,LO

ID2

ID2'

V

I

t

1/fLO

gLO(t)

VLO(t)

(a)

(b)

(a) (b)

Figure 3.10: (a) Topology of UWB subharmonic IRM (b) Diode current configura-tion

54

After ID1 and I ′D2 going through the IF directional coupler, outputs IF1 and IF2

can be expressed as

IF1 = ID1 + (−j)I ′D2 = ID1 + (j)m+1(−j)nID1 = (1 + (j)m+1(−j)n)ID1 (3.10)

IF2 = −jID1 + I ′D2 = −jID1 − (j)m(−j)nID1 = −(j + (j)m(−j)n)ID1 (3.11)

We now see what signal will going out when harmonic index when (m,n) equals

(1,-2)

IF1 = (1 + (j)2(−j)−2)ID1 = (1 + (−1)(−1))ID1 = 2ID1 (3.12)

IF2 = −(j + (j)1(−j)−2)ID1 = −(j + j(−1))ID1 = 0 (3.13)

this means that no signal component will show on IF2 when fIF = fRF − 2fLO.

When (m,n) equals (-1,2)

IF1 = (1 + (j)0(−j)2)ID1 = (1 + 1(−1))ID1 = 0 (3.14)

IF2 = −(j + (j)−1(−j)2)ID1 = −(j + (−j)(−1))ID1 = −2jID1 (3.15)

2fLO − fRF only appears on IF2. If the wanted signal is fIF = fRF − 2fLO, then

2fLO − fRF is the image signal. Therefore, the output signal will appear on IF1

and the image signal is totally eliminated by this proposed subharmonic IRM. As

the same results, if fIF = 2fLO − fRF is the wanted signal, the image signal is

eliminated and the output signal will appear on IF2.

The fundamental mixing signal appears when (m,n) equals (1,-1) or (-1,1). For

55

(m,n) equals (1,-1)

IF1 = (1 + (j)2(−j)−1)ID1 = (1 + (−1)j)ID1 = (1− j)ID1 (3.16)

IF2 = −(j + (j)1(−j)−1)ID1 = −(j + j(j))ID1 = (1− j)ID1 (3.17)

and (m,n) equals (-1,1)

IF1 = (1 + (j)0(−j)1)ID1 = (1− j)ID1 (3.18)

IF2 = −(j + (j)−1(−j)1)ID1 = −(j + (−j)(−j))ID1 = (1− j)ID1 (3.19)

The above two equations show that not only the fundamental mixing signal but

also the image signal reach output ports.

3.4.2 Circuit Realization and Measurements

Photograph and configuration of the proposed UWB subharmonic image rejection

mixer are shown in Fig. 3.11 and Fig. 3.12. MA-COM JHS-115, which is a

surface mount quadrature hybrid with bandwidth of 40-80MHz, is used as the IF

quadrature hybrid of the proposed mixer. The total circuit consists of a UWB

RF/LO directional coupler, two diodes, a low-pass filter, a high-pass filer, and an

IF quadrature hybrid.

Fig. 3.13 shows the conversion loss versus local power of the image rejection

mixer. The conversion loss is lower than 15dB when local power is larger than

8dBm. Fig. 3.14 is the graph of RF to IF conversion loss versus RF frequency,

conversion loss of image signal is also shown in this figure. The conversion loss of

RF to IF is lower than 20dB when RF frequency is from 3-13GHz, totally covered

the entire UWB bandwidth.

56

Figure 3.11: Photograph of the proposed UWB subharmonic IRM

57

RF/LO hybrid

Ground pad

LPFHPF

Bond wire

IF hybrid

RF

LOIF

IF

IF

IFLO

RF

Figure 3.12: UWB subharmonic IRM circuit configurationIF Frequency (MHz)

0 20 40 60 80 100 120

Imag

e R

ejec

tion

(dB

)

0

5

10

15

20

25

LO Power (dBm)

0 2 4 6 8 10 12 14 16

Con

vers

ion

Loss

(dB

)

10

15

20

25

30

35

40

45

IF Frequency (MHz)

0 20 40 60 80 100 120

Con

vers

ion

Loss

(dB

)

10

15

20

25

30

35

40

45

IFImage

3.16 3.17

3.13

Figure 3.13: Conversion loss versus local power of IRM

58

RF frequency (GHz)

2 4 6 8 10 12 14

Imag

e re

ject

ion

(dB

)

0

5

10

15

20

25

3.14 3.15

RF Frequency (GHz)

2 4 6 8 10 12 14

Con

vers

ion

Loss

(dB

)

10

15

20

25

30

35

40

45

IFImageCol 1 vs Col 17

Mean conversion loss = 16.5dB

Figure 3.14: Conversion loss versus RF frequency of IRM

Image rejection ratio is shown in Fig. 3.15. Most of the frequencies in UWB

bandwidth have the image rejection ration larger than 15dB, which is a basic spec-

ification of a wideband image rejection mixer.

Fig. 3.16 shows the IF bandwidth of the mixer. Conversion loss is measured

under the following conditions. LO frequency is fixed at 4GHz and varying RF

frequency from 8.01GHz to 8.11GHz. The mixer can have a better performance

from low IF frequency of about 40MHz to 80MHz. Fig. 3.17 is the image rejection

ratio versus IF frequency and Fig. 3.18 is the isolation of LO to RF versus RF

frequency.

59

RF frequency (GHz)

2 4 6 8 10 12 14

Imag

e re

ject

ion

(dB

)

0

5

10

15

20

25

RF Frequency (GHz)

2 4 6 8 10 12 14

Con

vers

ion

Loss

(dB

)

10

15

20

25

30

35

40

45

IFImage

3.14 3.15

Figure 3.15: Image rejection ratio versus RF frequency of IRM

IF Frequency (MHz)

0 20 40 60 80 100 120

Imag

e R

ejec

tion

(dB

)

0

5

10

15

20

25

LO Power (dBm)

0 2 4 6 8 10 12 14 16

Con

vers

ion

Loss

(dB

)

10

15

20

25

30

35

40

45

IF Frequency (MHz)

0 20 40 60 80 100 120

Con

vers

ion

Loss

(dB

)

10

15

20

25

30

35

40

45

IFImage

3.16 3.17

3.13

Figure 3.16: Conversion loss versus IF frequency of IRM

60

IF Frequency (MHz)

0 20 40 60 80 100 120

Imag

e R

ejec

tion

(dB

)

0

5

10

15

20

25

LO Power (dBm)

0 2 4 6 8 10 12 14 16

Con

vers

ion

Loss

(dB

)

10

15

20

25

30

35

40

45

IF Frequency (MHz)

0 20 40 60 80 100 120

Con

vers

ion

Loss

(dB

)

10

15

20

25

30

35

40

45

IFImage

3.16 3.17

3.13

Figure 3.17: Image rejection ratio versus IF frequency of IRMLocal Power (dBm)

0 2 4 6 8 10 12 14 16

Con

vers

ion

Loss

(dB

)

10

15

20

25

30

35

40

RF Frequency (GHz)

2 4 6 8 10 12 14

Isol

atio

n (d

B)

12

14

16

18

20

22

24

26

3.7 3.18

Figure 3.18: Isolation of LO to RF versus RF frequency of IRM

61

62

Chapter 4

Conclusion

63

In this thesis, a five-section 3-dB quadrature hybrid realized by conventional

PCB process with Rogers RO4003 has been successfully demonstrated. The pro-

posed modified VIP couplers have solved all three of the problems that a multi-

section 3-dB quadrature hybrid has always encountered, including an extremely

tight-coupled center section, equalizing modal phase velocities, and minimizing the

discontinuity effect between each section. VIP couplers with compensated stubs

have shown that return loss and isolation can be effectively improved. The proposed

directional coupler has shown a near-constant coupling over an ultra-wideband fre-

quency bandwidth of 1.5-13.5GHz, return loss and isolation are better than 13dB

over the entire bandwidth.

We have demonstrated an UWB subharmonic quadrature-IF mixer with con-

version loss better than 15dB when RF frequency is from 3-13GHz. The minimum

layout area and fewest numbers of elements are the benefits of this mixer. I/Q

amplitude deviation has shown fewer than 3dB in the bandwidth.

An UWB subharmonic image rejection mixer has also demonstrated in this

thesis. The mixer, when connected with an IF directional coupler, exhibits 15-

18dB conversion loss and 10-22dB of image rejection. IF bandwidth can cover the

frequency range about 40-80MHz due to the limitation of the IF directional coupler.

64

Bibliography

[1] V. Sedghi, A. Abdipour, and A. Mohammadi, ”A flat conversion gain subhar-

monically pumped image rejection mixer in mm-wave band” Microwave and

Millimeter Wave Technology, ICMMT 4th International Conference Aug. 2004,

pp. 312 - 315.

[2] P. Blount, ”An LMDS subharmonically pumped image reject mixer” GaAs IC

Symposium, 23rd Annual Technical Dig., pp. 41 - 44, Oct. 2001.

[3] E. G. Cristal, and L. Young, ”Theory and tTables of optimum symmetrical

TEM-mode coupled transmission line directional couplers,” IEEE Trans. Mi-

crow. Theory Tech., vol. MTT-13, pp. 544-558, Sept. 1965.

[4] S. Uystal and H. Aghvami, ”Synthesis,design, and construction of ultra-wide-

band nonuniform quadrature directional couplers in inhomogeneous media,”

IEEE Trans. Microw. Theory Tech., vol. MTT-37, no. 6, pp. 969-976, Jun.

1989.

[5] Duncan K. Y. Lau, Steve P. Marsh, L. E. Davis, and Robin Sloan, ”Simplified

design technique for high-performance microstrip multisection couplers,” IEEE

Trans. Microw. Theory Tech., vol. MTT-46, no. 12, pp. 2507-2513, Dec. 1998.

65

[6] R. M. Osmani, ”Synthesis of Lange couplers,” IEEE Trans. Microw. Theory

Tech., vol. MTT-29, no. 12, pp. 168-170, Feb. 1981.

[7] E.M. Bastida and N. Fanelli, ”Interdigitated coplanar directional couplers,”

Electron. Lett., vol. 16, no. 17, pp. 645-646, 1980.

[8] A. M. Pavio, and S. K. Sutton, ”A microstrip re-entrant mode quadrature

coupler for hybrid and monolithic circuit applications,” IEEE MTT-S Symp.,

Dig., pp. 573-576, May. 1990.

[9] L. Lavendol, and J. J. Taub, ”Re-entrant directional coupler using strip trans-

mission line,” IEEE Trans. Microw. Theory Tech., vol. MTT-13, pp. 700-701,

Sep. 1965.

[10] S. B. Cohn, ”The re-entrant cross section and wideband 3-dB hybrid couplers,”

IEEE Trans. Microw. Theory Tech., vol. MTT-11, pp. 254-258, Jul. 1963.

[11] Shelton et al., ”Tandem couplers and phase shifters for multi-octave band-

width,” Microwave, vol. 4, pp. 14-19, Apr. 1965.

[12] Y. Konishi, I. Awai, and Y. Fukuoka, ”Newly proposed vertically installed

planar circuit and its application,” IEEE Trans. Broadcast., vol. BC-33, no. 1,

pp. 1-7, Mar. 1987.

[13] Y. Konishi, I. Awai, Y. Fukuoka, and M. Nakajima, ”A directional coupler of

a vertically installed planar circuit structure,” IEEE Trans. Microw. Theory

Tech., vol. MTT-36, no. 6, pp. 1057-1063, Jun. 1988.

66

[14] H.C. Chen and C.Y. Chang, ”Modified vertically installed planar couplers for

ultrabroadband multisection quadrature hybrid,” IEEE Microwave and Wire-

less Components Letters, vol. 16, no. 8, pp. 446-448, Aug. 2006.

[15] C. Zhao and I. Awai, ”Applications of the finite difference techniques to

the compensated VIP 3dB directional coupler,” IEEE Trans. Microw. Theory

Tech., vol. MTT-44, no. 11, pp. 2045-2052, Nov. 1996.

67