Low Noise Photodiode Amplifier Write-Up

EECS 189 A/B

Professors:R. NelsonJ. Larue

Mentors:Allen Kine

George Horansky

Faruk Leric Jr. Brian Gorda Eugine Rubin

March 19, 2007

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Low-Noise Photodiode Simulation Writeup

With their low-input currents, FET input operational amplifiers are quite

commonly used in monitoring photodetectors. The most common of these configurations

are usually applied to photodiodes. The photodetector bridges the gap between a physical

phenomenon, such as light in our case, and the electronics used to detect and interpret

these phenomena. Selection of the appropriate amplifier architecture depends on ones

choice of noise, bandwidth, offset, and linearity. A traditional topology that is quite

commonly used in photodetection is as follows:

1

Here the photodiode is in series with the input of an operational amplifier where

ideally zero current flows. RIN and CIN are used to reduce DC and AC errors caused by

input bias currents. CIN shunts signal so that the noise inverting input approximates AC

ground. However, at the same time RIN also increases noise at the output by a factor of

1 Bonnie C. Baker, Comparison of Noise Performance Between FET Transimpedance Amplifier and A Switched Integrator, Burr Brown, 1993

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4KTRB times the noise gain of the circuit. The op-amp has feedback set by R2. This

resistance in a current-to-voltage converter such as the one presented above, largely

determines noise and bandwidth as well as gain. The spectral density of the resistor is

given by 4KTR, excluding non-ideal resistor behavior, and appears directly at the output

of a current-to-voltage converter. As we can see from the mathematical relationship,

increasing R not only increases output noise, but it also increases the output signal by a

direct proportionality. Consequently, signal-to-noise ratio tends to increase by a square-

root relationship. Noise from the operational amplifier also influences the output, with

an additional effect introduced by the high feedback resistance R2 as well as the

capacitance C2. Amplifier noise sources are usually modeled by the input noise current,

in, and the input noise voltage en shown as follows:

2

The current noise in flows through the feedback resistor R experiencing the same

gain as the signal current Ip. The DC noise voltage gain is given by 1+R/RD. This gain is

2 Photodiode Monitoring With Op Amps, Burr Brown, January, 1995

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kept small by the high diode resistance RD. However, at higher frequencies the above the

current bandwidth capacitance large values CD of the photodiode compared to the

feedback capacitor starts having a significant effect on the noise gain. This in turn

influences the gain of en in a proportional manner. Since the feedback resistance is

commonly made large, the effect can take place at relatively lower frequencies for small

values of C in parallel with the parasitic capacitance. The noise gain gradually rises and

eventually terminates at the intersection of given bandwidth curve. The feedback

capacitance C, shunts the feedback resistance R and results in a pole leveling response at

1+CD/C. For large area diodes, the capacitance CD can be in the picofarad range. This

could have a detrimental effect since the noise gain could be amplified by a factor of a

hundred as well. As such, gain peaking phenomena are a common side effect of high-

feedback op amp resistors and need to be carefully studied to maintain good behavioral

characteristics of the circuit. The effects of not properly selecting the feedback

capacitance can result in overshoot, response peaking, poor settling and quite possibly

oscillation. All of these effects are highly undesirable and need to be accounted for in the

architecture and design of the proper circuit topology.

Besides the DC and transient performance of the circuit, which is a function of the

overshoot, settling time and oscillation, we must also account for the noise characteristics

of the circuit. In understanding the current-to-voltage noise performance, it is important

to recognize that the signal current and noise voltage encounter different frequency

responses. The current-to-voltage is flat with frequency until the feedback impedance is

rolled off by the stray capacitance. The majority of the op amps bandwidth quite

commonly serves to amplify the noise error and not the signal. This is also typically the

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dominant source of noise for high feedback resistances. A typical noise versus feedback

resistance plot will usually behave in the following manner:

3

Upon inspection of the plot, we note that as the feedback resistance of a current-to-

voltage converter increases, the dominant noise source changes from the op amp to the

resistor and back to the op amp under gain peaking conditions. These effects are shown

for several op amps. Nameley, the OPA 404, the OPA 128 which we will utilize in our

design, and the OPA 111/OPA2111. The OPA 128 is a highly desirable op amp in our

case because it exhibits low noise characteristics in the middle range that is due to the

amplifiers low bandwidth that is 1 MHz (typically). It also excels in its very low DC

error characteristics as its input currents are a mere 0.075pA. Thus it is also a very low

input current device that we can take advantage of in our low noise amplifier design.

To understand the detrimental effects of noise even further, it is important to give

a brief overview of the governing principles behind low noise amplifier systems. The

first stage of amplifier design is the most prone of all to noise disturbance because noise

and signal are related at the input in an application. Noise when referred to the input is

3 Photodiode Monitoring With Op Amps, Burr Brown, January 15, 1995

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independent of amplifier gain or input impedance. The Noise Figure is defined as the

ratio of the signal to noise at the output to that of the input. This can be expressed as

follows:

NF= (s/n)I/(s/n)O,

where (s/n)I is the signal to noise ratio at the input, and (s/n)O is the signal to noise ratio at

the output of the device under test conditions. Hence, the noise figures can be referred to

the input. We know that noise can be modeled as a series of Gaussian distributions and

that uncorrelated noise combines in quadrature. Mathematically, this can be expressed as

follows:

e2tot = e21 + e22 + e23 +..+ i2nR2n

where e is a voltage input. This simple relation demonstrates that the total noise

accumulates as the sum of the squares of individual noise contributions. Consequently,

the success of the design will largely be influenced by the ability to curb and control

noise figures. To illustrate the significance of the phenomenon, we will proceed with a

relatively simple model as follows:

Here G1, G2, and G3 serve as hypothetical amplifiers of several stages in our circuit

topology. It is noteworthy to point out that it is impractical to put all the gain in one

location because of power, heating, material, and size related issues. To calculate the

signal and noise at the output the following is done:

4 EE Lectures, 2002

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Sout = G1G2G3Sin

This can be extended to an Nth order system for theoretical purposes. We then have the

following at the output:

(eout/Sout)2 = (e21 + e22/G1 + e23/(G21G22))/S2in

Therefore we can deduce from this relation that it is desirable to maximize gain at the

input stage. The first stage would boost the signal enough for the transmission down a

cable and should be large enough such that environmental noise is not significant enough

to affect the functionality of the circuit.

To control noise, it is imperative to recognize that gain peaking effects are the

primary noise limitation with a commonly configured high feedback resistance topology.

To limit gain peaking, additional capacitance is commonly added to bypass the

detrimental effects posed by the feedback resistor. Simply adding feedback capacitance

to the photodiode amplifier however, reduces the circuits high-frequency noise gain.

This does not actually reduce noise bandwidth, but reduced gain peaking decreases the

high-frequency effect of the op amps noise voltage. The amount of capacitance will vary

depending on the impedance posed by the resistor. Further, although adding feedback

capacitance is an effective means of reducing and controlling noise gain, it also decreases

signal bandwidth by the same factor. That bandwidth is already low with high feedback

resistance. Consequently, the end result could be a response of a kilohertz or less.

Circumventing the bandwidth problem is accomplished by utilizing a composite

amplifier. It consists of two op amps with the added one for phase compensation control.

A common topology may look as follows:

7

5With this composite structure, internal feedback controls the frequency response

of the gain added by the second op amp A2. At DC, we know that capacitors block the

feedback and this is the purpose of C1 in the schematic. The overall open-loop gain will

be given by the product of the two amplifiers. The gain will be rolled off by the open-

loop pole of A1 and by the response established by for A2 by C1 and R3. This

configuration is also a two pole roll off; therefore it is necessary to reduce it to establish

frequency stability and avoid oscillations. It is interesting to note that minimal noise, or

offset, is added by the second op amp A2 since this amplifier is preceded by the high gain

of A1. This topology is especially useful in situations where low level signals have

greater sensitivity to noise such as in our case. However, the feedback approach adds

simulation complexity to optimizing the design and little practical advantage over current

to voltage stage followed by a modified two pole filter stage.

Another design consideration that has to be accounted for is the bandwidth of the

circuit. This is especially important for current-to-voltage converters since the total

output noise increases in proportion to the square root of system bandwidth because a 5 Photodiode Monitoring With Op Amps, Burr Brown, January, 1995

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broad noise spectrum is encompassed. Typically there is also the added conflict of an

optimum signal-to-noise ratio and the bandwidth. This optimum occurs at a very high

gain. However, for current-to-voltage converters the situation is dire because they are

bandwidth limited. For instance, 0.5pF of stray capacitance around a 100M feedback

resistor pulls the signal bandwidth from a megahertz level to a kilohertz level. To

minimize this, know as stray shunting, low capacitance resistors and assembly

precautions are used. On a practical level, mounting feedback resistors on standoffs

reduces capacitive coupling with printed circuit boards. This is one possible source that

usually adds more capacitance to the overall circuitry. Bandwidth beyond parasitics such

as capacitance requires lower feedback resistance and accompanying lower converter

gain. Although it is very important to improve the bandwidth through voltage gain,

output noise increases which is undesirable. As such, a right compromise must be struck

for a given design.

The traditional transimpedance amplifier topology is preferred because it gives us

a real time representation of light excitation at the output of the amplifier. Alternative

topologies on the other hand, such as the switched integrator, give a time-averaged

representation of the input information from the photodetector. Despite the limits to the

solution in real time, such as bandwidth limits and settling time problems set by the

amplifier, this approach is optimal for low and medium bandwidth applications where

information about the amplitude and shape of the input signal is important such as in our

case. The switched integrator requires careful analysis of the frequency oscillation and

the effect of its characteristics as the switched integrator. Additionally, the switched

integrator topology is inherently limited in terms of bandwidth because of restrictions

imposed by the slew rate, settling times, and switching times of the integrator itself. The

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only advantage it has over the traditional topology according to the Burr-Brown article is

its relatively better noise performance.

Much of the discussion so far has dealt with the architectural design and topology

most advantageous for the low noise amplifier to be used in photodetection. Since

topology issues have been resolved, we are now ready to discuss concrete simulation

outcomes. The simulation software that we utilized for our purposes was Linear

Technologys (LTI) TINA Spice Simulation software. It is a user friendly software

available free for download from Texas Instruments web-page. The decision to use the

software is partly influenced by its ability to simulate noise figures that are not as easily

obtained in PSPICE, and the inclusion of OPA 277 and OPA 128 in simulations model

library. The requirements for the low noise amplifier are as follows: root mean square

circuit noise equivalent to one picoampere of photocurrent for the Hamamatsu S1336-44

BQ photodiode. The overall circuit response for a 100 Hz photocurrent square wave

varies dynamically from 1 pA to 25 mA. The output voltage for the specified

photocurrent must be roughly 10 Volts. For a 100 M feedback resistance the required

RMS noise would be 1 pA * 108 /2 or 70.7 V. The design will use rechargeable

Nickel metal hydride batteries that have a 9V case configuration . The primary circuit

operation will be between 16.6 and 14 volts. For simulation purposes the circuit analysis

will use plus and minus 15 volt supplies.

The amplifier schematic for the low noise amplifier is as follows:

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The schematic consists of a model photodiode which is composed of a diode shunt

resistance, a current source for the dark current, a current source for the photo current,

amplifier and a junction capacitance. The first op amp is an OPA 128KM. It is a

monolithic amplifier that is commonly used in photodetection schemes. A noise free

cascode and low-noise processing give it superb low-level signal processing. The choice

of this particular amplifier is also based on its ultra-low bias current of roughly 75fA

max, its low offset voltage, which is 500V max, its low noise currents, and low

bandwidth. The amplifier is also optimized for low power consumption purposes. This

will allow our circuitry to last longer for longer time durations and will require less

charging frequencies as light spectra are being studied. Its side terminals are tied to 15 V

voltage sources that are then grounded.

At the positive terminal of the OPA 128KM we have also modeled the parasitic

capacitance of the resistor. The feedback capacitance is the parasitic capacitance and the

capacitance required to suppress circuit oscillations. The output stage of the first op amp

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is followed by an RC filter as well as another amplifier. The second amplifier chosen for

photo detection purposes was the OPA277P. The choice of this op-amp was influenced

by the following operational characteristics: it has an ultra-low offset voltage of

roughly 10V, high open loop gain of 134dB, high common-mode rejection of 140 dB, and ultra-low bias currents. Asides from this, it also has a wide voltage supply range

spanning from 2 V up to 18 V. Its positive input polarity has been grounded while the

negative input is connected to the output of the first stage via a resistor. A modified 800

Hertz unity gain RC filter has been utilized for the purposes of noise control. The filter

was initially designed to be used with a higher bandwidth operational amplifier which

would have required additional filtering due to capacitive gain peaking of the first stage.

The filter is followed by the dual version of the OPA277 , that is the OPA2277, which is

used to create two unity gain differential signals which are 180 degrees out of phase. The

signals then go to a LT1167AC instrumentation amplifier with gains of either two or one

hundred. The conversion of the differential signal to a single phase signal by the

instrumentation amplifier results in a gain of two so that with the instrumentation

amplifier overall gain is 4 or two hundred. The output of the instrumentation amplifier

goes to two amplifiers which are used as RC filters and for gain. The filters were

designed so that their combined bandwidth was approximately 600 Hertz. The first filter

has gains of 1, 3, 5, 7, and 10. The second will has gains of 1, 10, 20, and 40. Following

the two filters there is a voltage follower that is basically used as load buffer. The three

amplifiers following the instrumentation amplifier are the OPA277P.

In light of the background information and our structural topology, it is only

befitting to continue the discussion of low noise amplifiers with simulation outcomes.

The simulations have been conducted in the following manner: capacitances C1, C2, and

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C3 have been varied in a manner to yield the most desirable transients, output and total

noise curves. The frequencies at which responses were noted are 100 and 400 Hz

respectively. Unless otherwise noted the photocurrent is one picoamperes, and a dark

current of 3 picoamperes. For comparative purposes, plots will be included in our

discussion to elucidate the subject matter. In the first attempt, we used the following

physical set of data:

Total Noise (uV)

Output Noise (uV/Hz) Cap. 1 Cap. 2

Frequency (Hz) Cap. 3

13.96 1.41 4p 4p 100 10n27.45 1.29 4p 4p 400 10n

The corresponding curves of the transient response, output noise and total noise looked as

follows:

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Transient Response

Output Noise

14

Total Noise

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From the table and plots we observe the following: the transient response is well

behaved. It is regular, periodic and experiences no oscillations, indicating that the circuit

is stable for the capacitance values listed. In a similar fashion, the output noise and total

noise experience flat responses with no peaks or dips. From the table we observe that the

output and total noise are 1.41 uV/Hz and 13.96 uV for 100 Hz respectively. For 400 Hz,

these figures are 1.29 uV/Hz and 27.45 uV. As such, we observe that at higher frequencies, the

output noise decreases while the total noise increases by a factor of roughly 2. If we perform the

same analytical procedure for the following capacitive parameters and frequencies we find that:

Total Noise (uV)

Output Noise (uV/Hz) Cap. 1 Cap. 2

Frequency (Hz) Cap. 3

13.99 1.41 1p 1p 100 10n28.44 1.43 1p 1p 400 10n

Transient Response

Output Noise

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Total Noise

Upon completion of the simulation with lowered capacitance, we find that the transient

response is well behaved once again. In a similar fashion, the output and total noise

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experience no dips or peaks. The respective responses indicate that gain peaking has not

become an issues as of yet. However at the same relative frequencies of 100 Hz and 400

Hz we find the following: at 100 Hz, the output and total noise are 1.41 uV/Hz and 13.99

uV respectively. Essentially the output noise has remained unchanged while the total noise has

increased by 0.03 uV. At 400 Hz, the output and total noise are 1.43 uV/Hz and 28.44 uV.

Therefore lowering capacitance has increased the output noise by .02 uV/Hz while the total

noise has increased by 0.99 uV. These changes indicate that lowering the capacitance makes

the circuit more prone to disturbance due to noise. As such, oscillations, which are highly

undesirable, are more likely to occur. This observation will be illustrated by the following results:

Total Noise (uV)

Output Noise (uV/Hz) Cap. 1 Cap. 2

Frequency (Hz) Cap. 3

13.99 1.42 10f 10f 100 10n28.5 1.44 10f 10f 400 10n

Transient Response

Output Noise

18

Total Noise

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After lowering the capacitance even further, we ultimately find that the circuit starts

oscillating. This is particularly obvious in the output noise plot, which shows an abrupt

peak around a frequency of 4kHz. Due to the high capacitance of the photodiode, the

amplifier becomes subject to instability and oscillation with lower feedback capactitance.

The perceived significance of this observation shows that controlling the capacitance

values is an important factor in determining whether the circuit topology will be well

behaved. Therefore we constrict ourselves to capacitance values in the picofarad range.

This observation will be illustrated by the following results:

Total Noise (uV)

Output Noise (uV/Hz) Cap. 1 Cap. 2

Frequency (Hz) Cap. 3

13.99 1.42 10f 10f 100 10n28.5 1.44 10f 10f 400 10n

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Transient Response:

Output Noise: Spot noise for 10fF capacitance

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Total Noise: 10fF capacitance

Circuit simulations for conditioning where oscillations begin after a step input are

difficult to simulate accurately as the software programs tend not to converge to a

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solution. The oscillation condition is better shown using phase plots. However, noise

plots also indicate the potential for oscillation.

This is particularly obvious in the output noise plot, which shows an abrupt peak

around a frequency of 4 kHz. The perceived significance of this observation shows that

controlling the capacitance values is an important factor in determining whether the

circuit topology will be well behaved and yet meet the bandwidth and transient response

requirements.

While simulations shown above used had a photo current input of 1 pA, the

following simulations use a photo current input of 1nA in order to match the bench test

results which will be discussed later. The output of the OPA128KM is 0.1 volts which is

increased to 0.4 V by the differential gain of two and the instrumentation amplifier gain

of two.

The circuit simulation circuit topology looks as follows:

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The amplifier bandwidth shows that our low noise amplifier will meet the bandwidth

specification. With a lower, but stable capacitance values the bandwidth increases. Very

low picofarad capacitors have higher percentage tolerance limits as well as less tolerance

for the effects of parasitic capacitances. For an effective one picofarad feedback

capacitance, the bandwidth is as follows:

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For bench testing we do not have the ability to verify the circuit response photo current

response with any accuracy to compare to the simulation results. However, we can

measure the circuit response to a voltage input by attaching a 50 meg Ohm resistor to the

non-inverting input which modifies the circuit input to a unity gain. The topology of this

circuit looks as follows, however the second schematic sheet is not shown as it is

unchanged from the photo current response shown above:

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The above figure contains the input waveform and the circuit response. However, the

input waveform has been multiplied by four to match the amplitude of the circuit

response.

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Asides from the aforementioned simulation responses, we also need to explore the

bandwidth response of the circuit. The simulations include the photodiode box and the

instrument amplifier in the amplifier box. For the OPA128KM, the time response has a

1pA current which yields 100uV. The two out of phase amplifiers add a gain of two.

Further, the instrument amplifier has a gain of 100 so that the output of the

instrumentation amplifier is 20 millivolts.

Amplifier Bandwidth Response

The amplifier bandwidth shows that our low noise amplifier will be a low bandwidth

circuit. This is one of the specifications that our circuit has to meet. Hence, this

requirement will be satisfied as shown. The bandwidth response will be restricted

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between roughly 200 and 400 Hz. It is also important to explore the amplifier time

circuit. The topology of this circuit is looks as follows:

In the amplifier box it is important to point out that the detector box and amplifier do not

have a common ground. Additionally, the instrument amplifier LT1167 gains are set by

the resistor between pins rg1 and rg2. R10 and R11 are selected by a switch and are not

actually in parallel. The actual time response itself of the circuit is given as such:

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From the simulation we deduce that the circuit is well behaved and experiences no gain

peaking which could potentially make the circuit unstable or oscillate.

Given the architecture discussion, it is also important to give a brief overview on

some of the practical issues on the concrete design level. The considerations to be

accounted for include external noise sources for instance. With its high resistance, a

current-to-voltage converter is fairly sensitive to noise coupling from electrostatic,

magnetic and radio frequency sources. As such, they will require great care. Care can be

exercised by means of shielding, proper grounding, and the physical location of the actual

components. Electrostatic coupling can emanate from the power line for instance in

practical design. Therefore voltage differences between objects are impressed on various

capacitors; consequently any voltage difference is coupled as noise current. To avoid this

commonly encountered phenomenon, shielding is commonly resorted to. Shielding

allows one to intercept the coupled current and shunt it to ground. However, it is

important to avoid parasitic capacitances that are created by the shields themselves.

Therefore, the shields must be returned to the signal common to avoid the coupling

effect. Therefore, this concludes our design approach analysis.

PCB Board Layout, Assembly, and Test

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PCB board layout design is an important aspect in the correct functionality of a

low noise amplifier. There are several different areas that need to be addressed when

laying out the components which are necessary for our circuit. Both layout design and

assembly are vital parts in successfully creating a functioning PCB board.

The layout portion of the PCB board is initially constructed with the computer

software program, ExpressPCB. Through this program the components we chose in the

design and simulation stage can be physically placed onto a PCB board. This is

accomplished by placing the necessary pins in the board which correspond to the correct

sizing of the component that is being placed on the board. Yellow silkscreen outlines the

body of the component, in-between the pins, which acts as a visual aid when laying out

and assembling the board. The name of the component can also be written in the

silkscreen to further aid in organization of the board and component placement. The

components we used for mounting on the PCB board are resistors, capacitors, operational

amplifiers, instrumentation amplifiers potentiometers as variable resistors, and a

photodiode.

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This picture represents what our final board layout looks like as a PCB file. Each

component is traced with the yellow silkscreen and labeled, and each label corresponds to

the component part used in the design/simulation.

Furthermore, we chose to use a PCB board with two independent layers. A two

layered board allowed us to use the top layer for component placement, and the bottom

layer was used as the common ground, and for connections that would have required

crossing lines on the top layer. Since we used only analog components we were able to

use a single common ground to connect all the components that required grounding.

These are pictures of our actual board after fabrication. The picture on the left is the top

layer of our board before the components were soldered on, and the pictured to the right

is of the bottom, or ground layer.

Another important consideration of layout design is spacing. Initially we

designed two separate boards which would be connect through a cable, however through

the design/learning process we found that combining our two boards into a single board

would reduce noise, and more importantly cost.

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This picture represents an early design where the photodiode socket , the photodiode

amplifier, the unity gain filter stage, the dual amplifier used to create a differential signal

are separate from the other portion of the circuit. Reducing the two boards into one did

reduce cost and also reduced the area, thus creating a more difficult task in laying out the

circuit. Placement of the components is important when considering noise. Running

connection wires next to other wires causes coupling and parasitic capacitance, therefore

the placements of each component needs to be located in an area which is within ideal

proximity to the other components connected to it.

In addition to spacing, the use of our more complex parts in ExpressPCB required

custom pin configurations to be created. The photodiode, the variable resistors and the

standoff required custom creation and placement of the pins of these components

required calculation down to the thousandth of an inch. While this task was not

overwhelmingly difficult it did require precision as many of our comment pins are closely

arrange and the correct placement of the pins is vital

for a successful PCB board design.

The photodiode also posed a unique challenge in the

layout process. Our first stage of amplification is

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extremely sensitive to noise created by leakage currents and the current noise generated

by the photodiode. The board specifications could not be obtained from the manufacturer

which created a design problem that needed to be addressed. Pico ampere leakage

currents between the OPA128KM input pins where the photodiode is connected, or Pico

ampere current leakages in the areas where the photo diode, high impedance resistors and

OPA128KM input pins join. Leakages could occur on the board surface between the PCB

material and the conformal coating, through the PCB board. Guard rings could mitigate

the surface problem however board cleanliness and solder flux residues would still be a

concern. Another consideration is the parasitic capacitance of the high impedance

resistors, and the parasitic capacitance of the resistors to ground which could limit the

design bandwidth.

To solve both the leakage, parasitic capacitance and cleanliness problems standoffs

would be used as support points for the high impedance resistors and also supporting the

connection of the photo diode and the inputs to the OPA128KM amplifier. One of the

selection criteria for the OP128 amplifier was its availability in a TO-92 case which

would allow its input pins to be bent and not be inserted in the amplifier socket, and

connect directly to the photo diode, which was ideal for our design.

The standoffs are a non-conducting material which allows the photo diode, high

impedance resistors, and amplifier inputs to be independent from the board, thus solving

both the leakage and capacitance problems. Also as the PCB board and parts are new

their connection could be achieved with minimal flux residue.

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Once the layout design was completed and the board was fabricated, assembly

posed the next challenge. The first difficulty we encountered in assembly after part

mounting but before the insertion of the photodiode or amplifiers was a layout error

which resulted in insufficient clearance between a routing hole and the ground plane.

One of our pins was not completely etched from ground; this caused a direct connection

between ground and the power supply, thus causing our entire board to be shorted out.

For time constraint reasons we decided to sacrifice one of our extra boards and

disconnect the power supply in sections of the board, therefore we were able to deduce in

which general area of the board the short was occurring. Once we discovered where the

faulty connection was, we carefully drilled the short out and soldered a jumper lead to

complete the circuit connection. Additionally the plus and minus supplies were

interchanged for one amplifier and to solve this connecting lines were severed with a

small tool and jumper wires were installed.

Another assembly conflict we encountered was flux. In some area there was a

build up of flux and residue. The flux residue could create undesired conditions on our

board which caused our board to function incorrectly. The flux was not a problem in the

highly critical areas of the PCB. To remove the flux from our PCB board we cleaned the

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This is a picture of how the op-amp should be placed on standoffs and guarded, to avoid the leakage and grounding problems

surface with an alcohol solution which removed the flux sufficiently. The close

proximity of the components on the board also caused some assembly disagreement.

Close attention needed to be paid to ensure each pin was correctly connected to the

proper lead of its component. The clearance between the hole area to solder a lead and the

surrounding ground plane on the back of the PCB was minimal. The layout used the

manufacturers standard clearance which we in many cases should have enlarged. In

addition to that, the leads of each component needed to be isolated from the other leads

surrounding each component, and with some components confined to a small area contact

between the leads was difficult to avoid. Only one small surface mount integrated circuit

part type was used in the design and it was soldered using a binocular microscope. To

make sure each lead had clearance from another lead or the ground plane board was often

closely inspected. Shielding the PCB board from outside noise, such as electrostatic and

magnetic pickup was another important assembly consideration. To ensure our circuit

maintains a low noise signal exposure to outside interference was necessary in producing

a low noise signal. To account for outside pickup a fully enclosed box was purchased to

encase the board, and to mount switches and other connections. The top, bottom, and two

sides of the box were 0.050 inch steel and two sides for mounting switches and other

components was .060 inch aluminum. The box would be the primary shielding from any

electrostatic or magnetic pickup, with any required additional shielding being

experimentally determined. The box did offer sufficient shield, however the photodiode

requires exposure to light, and thus to accommodate the diode a hole was drilled through

the box which allowed light exposure to the surface of the photodiode.

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The hole necessary for the diode allowed a strong signal to be created from the

photodiode, however the hole also exposed the circuit to noise (pickup). To allow the

photodiode light, while shielding the rest of the board from noise a non-reflective steel

tube was placed through the hole in the box. The photodiode was placed in one end of

the tube while the other end was open to accept light for the diode. This offered a partial

solution to our noise concern. However, it is anticipated that there will have to be a

conducting mesh screen over the end of the tube. The design of the box and PCB did not

have any common ground points. The battery power is contained within the box. A

problem arose with the addition of the tube, and when the photo diode was inserted into

the tube. The photo diode had little response to light. The cathode of the photo diode is

connected to the TO-92 case, and the cathode is also connected to the inverting pin of the

OPA128KM. The placement of the photodiode in the tube created a connection between

the diode and the box, which caused our circuit to function improperly. After some

experimentation it was determined that the isolation between the box and the PCB ground

was approximately 30 meg ohms. Since the photo diode shunt impedance is typically 600

meg ohms the circuit was essentially converted to a voltage amplifier and not a trans-

impedance amplifier. To avoid the connection between the diode and the box, an insulting

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Picture of the tube in which the photodiode fits into for shielding. The tube is connected to the box

dielectric was placed between the photodiode and the tube to avoid the electrical

connection.

The testing portion of our project was an important area which allowed us to

make the correct modifications to our circuit. Since we encountered several problems

during assembly testing our circuit was necessary in determining exactly where a short,

oscillation, improper connection occurred. We found how close our measured gain

values were in comparison to the theoretical values.

The error percentage of the measured value in comparison to the theoretical value is very

small, which is important for our circuit to function properly. In addition to measuring

the gain and voltages across the components of our circuit we also measured the output

noise as a Gaussian distribution.

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The output noise of 52.8 micro volts is well within our ceiling of 70 micro volts of output

noise. Furthermore, we also took the FFT for the fast Fourier transform of the noise,

which gave us our frequency response of the noise.

The test confirm that our circuit is overall well behaved and within our low noise

specification. Testing was a vital portion of our project which allowed us to both adjust

our circuit in order to obtain our desired results, and confirm the noise and gain values

were within the appropriate range.

In conclusion our project was a success. We completed our project on time with

all parts functioning correctly. In addition to that we met all of our initial design

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Picture of the noise at the output with a gain of 1600.

specifications which were set by our professors. The output noise signal, as a function of

the input, measured to be 50 micro volts which is under our maximum of 70 micro volts.

The numerous gain settings on our box all produce and output gain that is very near the

expected value. Through this project we learned several valuable lessons, which pertain

to not only the technical side of engineering, but also the teamwork and communication

aspect of engineering. Our group learned to work together and communicate to solve

problems and meet deadlines. This project has taught us real life engineering skills which

we will be able to demonstrate as we make the transition from student to employee.

The future advice we would like to extend to next years low noise photodiode

amplifier group is, first, design is the most crucial step. Make sure your design is correct

as even small mistakes in the design take much more time to solve and correct once the

board is fabricated. In addition to that, time management is another important topic for

this project, and creating a time schedule in the first quarter would be a good idea.

Lastly, make sure everyone communicates and is assigned specific areas. Our group

successfully completed our project because we divided the workload, and made sure to

communicate and work together as a team.

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Appendix

Software (Board Design): Express PCB

Software (Simulations): TINA PSPICE

Parts purchased www.digikey.com www.newark.com

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