chapter8 emc

Elec3017:Electrical Engineering Design

Chapter 8: ElectromagneticCompatibility

A/Prof D. S. Taubman

August 29, 2006

1 Purpose of this ChapterElectromagnetic compatibility refers to the ability of an electronic product tooperate correctly within its environment. There are two aspects to this: 1) theproduct must be able to operate correctly even in the presence of electromagneticinterference (EMI) from other electrical appliances; and 2) the product must notitself generate undue electromagnetic interference.The first aspect addresses the fact that EMI is unavoidable, so that electronic

products must be designed in such a way that they do not fall victim to itseffects. We take this aspect up first in Section 2. The second aspect addressesthe fact that all electronic products are sources of EMI. Inserting products whichproduce undue levels of EMI into the environment may make it impossible forother products to function. For this reason, acceptable levels of EMI generationare the subject of regulatory standards, rendering the sale of non-conformingproducts illegal. We consider methods to minimize generated EMI in Section4. As we shall see, many of the methods which reduce susceptibility to EMI(taking the perspective of a victim) also reduce the amount of generated EMI(taking the perspective of a source).Most likely, you will approach this subject with the view that electromag-

netic interference consists of unwanted RF emissions. Certainly this is true,but EMI occurs also at very low frequencies, right down to DC. At very highfrequencies, EMI may be radiated wirelessly, while at low frequencies we willbe more interested in EMI which is conducted through wires. This includes themains wiring which is shared by appliances in your home.Failure to pay proper attention to electromagnetic compatibility may have

serious adverse consequences. Here are some of the potential consequences, inorder of increasing seriousness:

• Poor or lost reception in wireless applications;

1

c°Taubman, 2006 ELEC3017: Electromagnetic Compatibility

• Corruption of analog or digital data on transmission lines;• Malfunction of medical electronic equipment;• Malfunction of automotive microprocessor control systems; or• Inadvertent detonation of explosive devices.Since electromagnetic interference is a complex, multi-faceted topic, with

this brief chapter we cannot hope to make you an expert. Our treatment willfocus mostly upon basic principles of good design. We begin in Section 2 witha discussion of basic principles to minimize your susceptibility to EMI. This isfollowed by Section 3, which is concerned exclusively with the problem of com-municating information over transmission lines. Transmission lines effectivelyextend our electronic circuits over quite some distance, making them particu-larly vulnerable to EMI. The perspective in both of these sections is that EMIhappens and all we can do is minimize our susceptibility to it. In Section 4, welook at the techniques to minimize the amount of EMI which you produce.

2 Methods to Avoid Being a Victim of EMIIt is important to realize that EMI occurs not only between electronic prod-ucts, but also within an electronic product. That is, circuit elements from onepart of the product interfere with other parts of the product. The materialin the present section is concerned with both internal and external EMI. Thematerial is organized into three sub-sections, which can be classified as primary,secondary and tertiary levels of control.

2.1 Primary control: components and Layout

There are three types of electromagnetic coupling with which we are principallyconcerned at the level of circuit layout. These are covered under the followingheadings.

2.1.1 Inductive coupling

We use this term to refer to the susceptibility of conductive loops in your cir-cuit to alternating magnetic fields. The situation is illustrated in Figure 1.The alternating magnetic fields in question may be generated by nearby cir-cuit elements within the same product, or they may be due to propagating RFelectromagnetic waves from more distant sources1. In either case, an inducedelectromotive force (EMF) E = dΦ

dt is generated in the circuit loop, where themagnetic flux Φ is proportional to the total area contained within the loop. Theprimary defense, therefore, against inductive coupling is to keep sensitive signalconductors as straight as possible. This can be more difficult than you might

1At a fundamental level, there is no difference between these two sources, since uncontainedalternating magnetic fields always give rise to propagating EM waves.


circuit loop

enclosedflux, Φ

circuit loop

enclosedflux, Φ

circuit loop

enclosedflux, Φ

Figure 1: Inductive coupling: alternating magnetic fields create an EMF incircuit loops. These may be difficult to avoid due to layout constraints. Thefigure shows two IC’s with interconnected pins. These might be opamps with anoutput from one opamp connected to multiple inputs on other opamps.

Separation, d

Overlapping area, AY

XSeparation, d

Overlapping area, AY

X

Figure 2: Capacitive coupling between conductors X and Y.

think, particularly when signal lines are used to connect multiple components,as suggested by Figure 1. It is also worth noting that electronic componentsthemselves are part of the circuit. Their packages and leads may well formor contribute to circuit loops, which are susceptible to inductive coupling. Ofcourse, the worst offenders are likely to be inductors and transformers.

2.1.2 Capacitive coupling

Capacitive coupling can occur between any two closely spaced conductors, Xand Y , as shown in Figure 2. The capacitance formed between such conductors isroughly proportional to A

d , where A is the shared surface area of the conductorsand d is their separation. In practice, of course, both the area and separationmay vary from point to point, so this formula serves only as a guide to help youminimize capacitive coupling. Evidently, the two most obvious actions you cantake are: 1) minimize the surface area of either or both conductors (i.e., use thin,short wires); and 2) keep sensitive circuit traces as far away from interferencesources as possible.An important additional mitigation strategy is to insert a grounded conduc-

tor G, between X and Y . As shown in Figure 3, this replaces the capacitancebetween X and Y with two capacitances, one between X and G and anotherbetween Y and G. So long as G has low impedance and is tied to a stable fixedreference voltage, capacitive currents flowing between Y and G will have very


Y

XGY

XG

Figure 3: Decoupling conductors X and Y by inserting an intermediate groundconductor G. Note the effective capacitances formed between X and G and Yand G.

little impact on the potential of the ground conductor and hence the capaci-tive currents flowing between X and G. Similarly, capacitive currents flowingbetween X and G will have negligible impact on Y .There are various ways to implement this strategy in practice. If conductors

X and Y are running parallel to each other on a single side of a printed circuitboard (PCB), you may insert an extra ground trace between them. If conductorsX and Y are on opposite sides of a PCB, the ground conductor G might runin an internal PCB layer, of a multi-layered PCB — in the extreme case, thismight be an expansive sheet of copper, known as a ground plane. If one or bothof X and Y is a flexible conductor whose position is hard to fix, the groundconductor G might be a copper mesh which completely encloses one of X or Y— this is known as a shield.

2.1.3 Conductive coupling and ground planes

One particularly insidious form of electromagnetic coupling occurs when differ-ent parts of the circuit share conductive paths. We restrict our attention hereto the case where the shared paths correspond to ground or power lines, sincethese are the places where problems are most likely to occur.Figure 4 illustrates the problem of shared power lines. Since the power con-

ductors have non-negligible impedance, the current consumption behaviour ofdownstream components introduces voltage drops along both power and groundlines, which are experienced by upstream components. One way to mitigate thiseffect is to use conductors of large cross-sectional area, and hence very low resis-tance, for power lines. Unfortunately, this is of limited effectiveness in suppress-ing very high frequency power line transients. The problem is that the powerlines also have appreciable self-inductance which converts current transients intovoltage drops, according to

V = LdI

dt.

The self inductance of a conductor does not reduce substantially as cross-sectional area is increased.For this reason, it is important that you include local capacitors across the

supply rails of components which may consume large or rapidly changing cur-


Vcc (+5V)

Gnd (0V)

Vee (-5V)

opamp TTL LogicTTL Buffer

Vcc (+5V)

Gnd (0V)

Vee (-5V)


Figure 4: Power and ground lines shared by multiple circuit components, showingsupply currents. Of course, there will be additional currents flowing throughsignal lines, not shown here.

Vcc (+5V)

Gnd (0V)

Vee (-5V)


Vcc (+5V)

Gnd (0V)

Vee (-5V)


Figure 5: Power supply decoupling capacitors.

rents. This is illustrated in Figure 5. These capacitors are commonly known asdecoupling capacitors, or bypass capacitors. Typical values range from 1 nF to100 nF, depending on the application. For good decoupling of high frequencypower supply transients, ceramic capacitors are generally recommended. Spe-cial ceramic decoupling capacitors known asmonolithic capacitors are commonlyavailable, with typical values of 22 nF and 100 nF.Even with decoupling capacitors and low resistance power and ground lines,

appreciable levels of coupling can still occur. Further decoupling can be achievedby distributing power to your components via a star network, as shown in Fig-ure 6. In this case, separate circuit components do not share the same powerconductors. In practice, this can be quite difficult to arrange for each individualcircuit component. Moreover, capacitive coupling of different branches in thestar becomes increasingly likely as the complexity of the power distribution net-work grows. For these reasons, an intermediate solution is normally adopted, inwhich sensitive analog components in the circuit sit on one branch in the powerdistribution network, with noisy digital components powered from a separatebranch. Such a partitioned system is shown in Figure 7.


Vcc (+5V)

Gnd (0V)

Vee (-5V)

opamp

TTL Logic

TTL Buffer

Vcc (+5V)

Gnd (0V)

Vee (-5V)

opampopamp

TTL Logic

TTL BufferTTL Buffer

Figure 6: Star configuration for distributing power without shared conductors.

analogpartition

digitalpartition

Vcc (+5V) Gnd (0V)

Vee (-5V)

opamp

TTL LogicTTL Buffer

opampanalogpartition

digitalpartition

Vcc (+5V) Gnd (0V)

Vee (-5V)

opamp

TTL LogicTTL Buffer

opamp

Figure 7: Partitioned power and ground system.


The ground reference conductor plays a special role in most circuits, as acommon reference potential shared by many signals. Voltage drops along theground conductor may appear to the extent that it is used to carry transientpower supply currents, as discussed above. Voltage drops may also be causedby larger signal currents. These cannot be mitigated by decoupling capacitors,since signal line voltages are not generally supposed to be constant. Star andpartitioned ground networks are widely used to combat conducted coupling viaground lines.Another commonly employed technique is the use of ground planes. A ground

plane is an expansive sheet of copper, which serves to establish a solid groundreference potential. The use of ground planes can add to the cost of a printedcircuit board, since they typically consume an entire layer of the board. Onthe other hand, ground planes can provide very low resistance. Interestingly,ground planes can also lower the inductance experienced by ground currents.One reason for this is that high frequency ground currents will tend to find thepath through the plane which encloses the smallest possible loop area, leadingto lower inductance than is usually achieved by running separate ground tracksacross a PCB.It is tempting to think that a ground plane will cure all your ground-based

signal coupling problems; this kind of thinking, however, can get you into trou-ble. The main problem is that ground planes deprive you of control over thepaths which are actually taken by ground currents. This means that high tran-sient currents produced by line drivers or high speed digital IC’s may intersectwith the ground return paths taken by sensitive analog signals. Since currentsfan out within the ground plane in a complicated fashion, interference is difficultto avoid and may lead to quite unexpected coupling in high gain analog circuits.With this in mind, it is best to partition your ground plane into physically sep-arated regions: one for sensitive analog components; one for less sensitive linedriving components2; and one for switching components such as digital IC’s.These partitioned ground regions can then be connected via a star configura-tion, as suggested in Figure 7.

2.2 Secondary control: interface filtering

In Section 3, we take up the problem of reliable communication over transmissionlines. At this point, however, we note that the signals arriving at input interfacesto (or within) your product will generally be corrupted by noise and interference.These can be substantially suppressed through the use of input filters, whosepurpose is to remove frequency content which lies outside the range of thesignals you are actually expecting to receive. Without input filtering, noise andinterference power may substantially exceed the power of the intended signalthat you are trying to recover. Simple R-C filters are commonly employed,but more complex multi-pole filters might also be required. If the frequencies

2This one is optional. Many designs involve only two ground regimes: one for sensitivesignals and one for less sensitive signals with higher slew rates.


Π filterLM7805in out

ref

GND

VCC

SUPPLY -

SUPPLY +

regu

late

dou

tput

unre

gula

ted

inpu

t

Π filterLM7805in out

ref

GND

VCC

SUPPLY -

SUPPLY +

regu

late

dou

tput

unre

gula

ted

inpu

t

Figure 8: Filtering at the output of a regulated power supply to suppress EMI.

involved are not too high (e.g., less than 1 MHz), good input filters can beconstructed with the aid of opamps, so that the only passive elements requiredare resistors and capacitors. At very high frequencies, tuned circuits containingboth capacitors and inductors are generally required.An important source of electromagnetic interference is through mains power

lines. Common sources of EMI on mains power lines include:

• switches;• appliances with commutated motors (e.g., electric drills, kitchen tools,vacuum cleaners, etc.);

• electric light dimmers (these introduce strong switching transients partway through the 50 Hz mains cycle); and

• communication devices which use the mains wiring as a communicationsmedium.

Interference from such sources can be coupled into your product via its powersupply system. In fact, this can be a significant problem for audio amplificationand mixing equipment. Again, the solution is to filter out unwanted frequencycomponents, either before or within the regulated power supply system. The useof resistors and/or active amplifiers (e.g., opamps) in such filters is not generallyappropriate, since these would waste significant amounts of power. For thisreason, high quality power supply filtering generally involves both capacitorsand inductors. A common configuration is the Π filter shown in Figure 8.

2.3 Tertiary control: shielding as a last resort

High frequency radiated EMI can be strongly attenuated by enclosing yourelectronic product (or sensitive sub-systems) in metal shielding. To appreciatethe effectiveness of this as a solution, it is important to understand the conceptof skin depth. EM waves with frequency f , propagating through a metal withconductivity σ, have their amplitude attenuated according to

A (z) = A0e− zδ(f,σ) (1)


time

number of availabledefense deasures

cost of defensedeasures

time

number of availabledefense deasures

cost of defensedeasures

Figure 9: Availability and cost of EMI defense measures as the product designprocess evolves.

Here, z denotes the propagation distance and

δ(f, σ) =

sε0 · c2σ · π · f (2)

is the skin depth of the metal. At a distance δ (f, σ) into the surface of themetal, the amplitude is already reduced by 1

e , meaning that the RF power isreduced by e−2; at two skin depths, the amplitude and power are reduced bye−2 and e−4, respectively; and so forth. For effective shielding, two or threeskin depths may be sufficient.For copper, the ratio σ/ε0 (conductivity divided by the permittivity of free

space) is 6.5× 1018 s−1. Noting that the speed of light is c = 2.998× 108 m/s,we find that the skin-depth is given by

δcopper (f) =6.634 4× 10−2 m ·√Hz√

f

At 1 MHz, for example, the skin depth is a mere 66 μm. On the other hand, at100 Hz, the skin depth is 6.6 mm, which is a very thick sheet of copper indeed.Evidently, shielding cannot generally protect you from low frequency EMI.

Shielding also adds substantial cost and weight to your product, so it shouldbe seen as a course of last resort. Shielding is relatively easy to add as anafter-thought, if your design proves overly susceptible to EMI. Other, less costlymeasures, however, must be incorporated in earlier stages of the design process.This relationship between time, cost and available EMI defense mechanisms isdepicted in Figure 9.

3 Transmission Lines and Differential SignallingIn this section we devote special attention to the case of transmission lines.We use the term transmission line to refer to any wired signalling system for


communicating information over distance. This includes wires which link sep-arate electronic products (e.g., the cabling from a guitar pre-amp to a highpower audio amplifier), as well as signal lines which are used to connect distinctsub-systems within a single electronic product. What makes transmission linesparticularly important is that they extend over a significant distance, so theyare likely to be exposed to substantial levels of EMI. For this reason, we willneed to develop circuit solutions to minimize the impact of this EMI. Just whatconstitutes a significant distance depends upon the application at hand: thefrequencies involved; the sensitivity of the signals; and the expected levels ofelectromagnetic interference.We begin our discussion of transmission lines by considering the various

modes by which EMI may be coupled into a transmission line consisting of twoconductors: an active signal conductor; and a ground conductor. In this con-text, we show how ground shields can be beneficial when used with appropriatereceiving circuitry. We then turn our attention to differential signalling schemes,involving two active signal lines.

3.1 Single-ended signalling

Figure 10 illustrates an overly simplistic signalling environment, in which thetransmission line consists of a ground conductor and a single active signal line.The information (analog or digital) is communicated via the voltage differencebetween the active signal line (shown on top) and the ground potential (shownon the bottom). For simplicity, we think of the transmitter and receiver asphysically distinct electronic products and consider the various ways in whichEMI may find its way into the received signal. The most important of these areas follows:

Direct inductive coupling: This occurs in the presence of alternating mag-netic fields, such as those produced by propagating EM waves. The al-ternating flux Φ1 enclosed by the signal and ground conductors in thetransmission line (see Figure 10) produces an induced EMF, which is su-perimposed on the desired signal at the receiver.

Capacitive coupling to earth: This occurs when either of the conductors inthe transmission line are sufficiently close to another conducting surfacewhose voltage potential is a source of interference. We use the genericterm earth here to refer to this conducting surface. In practice, the closeinterfering surface might be the metal case of another electronic product.Alternatively, it might be a human being, whose body potential is oscil-lating in response to EM waves (as an antenna) or in response to anotherinterference source, to which it is capacitively or directly coupled. Ca-pacitive coupling can occur only if the potential of the nearby interferingsurface oscillates rapidly, or when the separation between the transmissionline and the interfering surface itself oscillates rapidly3.

3A time varying displacement between conductors produce a time varying capacitance,


Conductive coupling via earth: A potentially very serious EMI couplingmodality occurs if transmitter and receiver are both connected to themains earth wiring. Many electrical appliances have a metal case which isdirectly connected to the mains earth as a safety precaution. In the eventof an electrical fault in the appliance, a large current will flow to earth,causing a fuse or circuit breaker to open in the local mains breaker box.This topic is covered more extensively in Chapter 9. All that need be ap-preciated here is that electronic products which consume significant powerare usually earthed. Internal to the transmitter and receiver, the groundreference potential is also either directly connected to the earthed case(and hence to the external mains earth) or else there is strong capacitivecoupling to the case and earth (e.g., through the windings of a transformerused for power supply isolation). These connections are shown as dashedlines in Figure 10. They provide a mechanism for the coupling of earthcurrents into the signal recovered by the receiver. To see this, note thatearth currents produce potential differences between different points inthe mains earth wiring. These potential differences produce currents inthe transmission line’s ground conductor, which has non-zero resistanceand self inductance. In this way, earth currents produce end-to-end po-tential differences in the transmission line’s ground conductor, which aresuperimposed on the signal recovered by the receiver.

Inductive coupling via earth: Even if no other appliances share the mainsearth wiring path between the transmitter and the receiver, it is still pos-sible that earth currents appear, producing superimposed waveforms onthe received signal. A major cause of this is the coupling of alternatingmagnetic fields into the circuit formed by the transmission line’s groundconductor and the earth. This is identified in Figure 10 by the magneticflux term Φ2. Even though this is a less direct path for EMI coupling thanthat discussed above due to flux term Φ1, the area enclosed between thetransmission line and the mains earth wiring is usually much greater thanthat enclosed between the two conductors in the transmission line itself.

Having examined some of the more significant modes by which EMI maybe coupled into a single-ended transmission line, we now consider methods forminimizing its impact. The first measure we will consider is the use of shieldedcabling. There are still only two conductors in the transmission line, but one ofthese (the ground conductor) consists of a flexible wire mesh which completelysurrounds the inner conductor (the active signal line). This is illustrated inFigure 11.Shielding can substantially eliminate direct inductive coupling. To see this,

observe that EMF contributions E1 and E2 depicted in Figure 11 should bealmost identical, assuming that the magnetic field strength varies little betweenthe top and bottom edges of the cable. These EMF contributions produce

whose stored charge induces a time varying voltage. This mode of coupling, however, is rarelysignificant.


receivertransmission line

transmitter

earthearth currents

dtd /1Φ

dtd /2Φ

receivertransmission line

transmitter

earthearth currents

dtd /1Φ

dtd /2Φ

Figure 10: Coupling of electromagnetic interference to a transmission line in-volving single-ended signalling.

dtd /Φ

dtd /Φ

1ε2ε

dtd /Φ

dtd /Φ

1ε2ε

Figure 11: Shielded cable with one active signal line.

opposite superimposed waveforms on the received signal, so that the inductivecoupling contributions largely cancel.Shielded cables also largely eliminate capacitive coupling between external

interference potentials (or earth) and the inner conductor. Virtually all ca-pacitively coupled interference currents flow in the grounded shield, whose im-pedance is much lower than that of the inner active signal line. Such capacitivecurrents can still produce potential differences along the shield conductor, buttheir impact is generally much smaller than that of capacitive coupling to theactive signal line, especially considering the non-zero output impedance of thetransmitter’s line driving circuitry.Apart from providing a low impedance ground conductor, shielded cables do

not inherently provide a solution to the third and fourth EMI coupling modesdescribed above. To address these, it is necessary to break the ground-earthcircuit. This means that you should not tie the shield to both the transmit-ter’s ground reference point and the receiver’s ground reference point. To avoidambiguity, the convention is to tie only the shield to ground only at the trans-mitting end, allowing the receiver’s shield potential to float relative to its groundreference. The receiver then implements a differential amplifier which is sensi-tive only to the difference between the shield and the active signal line. Thissituation is depicted in Figure 12.As shown in the figure, the shield may still be tied loosely to the receiver’s

ground via a resistor, whose impedance is large compared to that of the shieldconductor. This is important if no other DC path exists between the transmit-


transmitterreceiver

opamp

R1R2

VO

R1

R2

RG

Vmax

Vmin

transmitterreceiver

opamp

R1R2

VO

R1

R2

RG

Vmax

Vmin

Figure 12: Single-ended transmission line with differential receiver.

ter and receiver’s ground reference point, since the receiver’s differential inputamplifier can only operate over a limited range of input voltages. Interferencewaveforms produced by earth currents, and inductive coupling via earth, appearacross this resistor instead of the transmission line’s shield conductor. So longas the voltages appearing across the resistor do not exceed the common modeoperating range of the receiver’s differential input amplifier, all will be well. Thefigure also shows how diode protection may be included to prevent the shieldvoltage from exceeding the safe operating range of the receiver’s differentialinput amplifier.Before concluding this sub-section, it is worth noting that a twisted pair can

serve as an inexpensive alternative to shielded cables. As the name suggests, atwisted pair is just a pair of separately insulated wires which have been twistedtogether. Twisting the wires together minimizes inductive coupling effects, sincethe magnetic flux enclosed by one twist is roughly cancelled by one twist isroughly cancelled by that enclosed by the next twist, as we move along thelength of the cable. Again, loose coupling of the ground wire at the receiver,together with differential amplification, minimize the impact of earth currentsand inductive coupling via earth. Twisted pairs, however, remain much moresusceptible than shielded cables to capacitive coupling of interfering surfacepotentials.

3.2 Differential signalling

While the EMI mitigation methods discussed in the previous sub-section can bequite successful, single-ended signalling strategies still suffer from the drawbackthat shield currents flow through one of the two signal conductors (the groundedshield). These shield currents include the currents produced by capacitivelycoupled interfering surface potentials. They also include the residual currentsproduced by mains earth noise, which cannot be completely eliminated sincethe shield must be tied at least loosely to ground at the receiver — in Figure 12,this is done by resistor RG and the two common mode voltage limiting diodes.Since the shield is one of the two information bearing conductors, the voltagedrop produced by shield currents will be superimposed on the received signal.The solution to this problem is to use two active signal lines, enclosed by a


transmitterreceiver

opamp

R1R2

VO

R1

R2

RG

Vmax

Vmin

transmitterreceiver

opamp

R1R2

VO

R1

R2

RG

Vmax

Vmin

Figure 13: Transmission line with two active signal lines, enclosed by a separategrounded shield.

transmitterreceiver

opamp

R1R2

VO

R1

R2

RG

Vmax

Vmin

transmitterreceiver

opamp

R1R2

VO

R1

R2

RG

Vmax

Vmin

Figure 14: Similar to Figure 13, except that the two active signal lines aredriven with voltage waveforms v (t) and −v (t), respectively. The transmitteressentially has two output drivers, one for each signal line, each with the sameoutput impedance. The sum of the two signal line potentials is equal to thetransmitter’s ground potential.

separate shield. One such configuration is illustrated in Figure 13. Another isdepicted in Figure 14. Both configurations offer excellent immunity to EMI. Thesecond involves a more complex transmitter which drives one signal line withthe opposite polarity to the other. One advantage of this is that the drivingimpedances of both signal lines should be identical, so that they are affectedroughly equally by residual EMI coupling. A second advantage of the differentialsignalling strategy in Figure 14 is that any EMI radiated by one active signalline is roughly cancelled by the EMI radiated by the other active signal line —see Section 4.With either of the configurations shown in Figures 13 and 14, the main

modes of potential EMI coupling are as follows:

Differential inductive coupling: This occurs in the presence of alternatingmagnetic fields, when the alternating magnetic flux enclosed between thetwo active signal lines generates an EMF which is superimposed on thereceived signal. Fortunately, the presence of a separate shield significantlyreduces the levels of high frequency EM radiation to which the activesignal lines are subjected, in accordance with equations (1) and (2). Tofurther mitigate against differential inductive coupling, the active signal


lines can be twisted together so that the EMF’s induced in alternate twistshave opposing amplitudes.

Impedance and length mismatches: In principle, EMI should affect bothof the active signal lines identically, so that the difference between theirsignal voltages remains unaffected. This is the reason for including a differ-ential amplifier at the receiver. One example which we have not yet men-tioned is antenna mode coupling. When suitably aligned, the alternatingelectric fields associated with propagating EM waves induce longitudinalcurrents in both of the active signal lines, acting as antennae. If the twoactive signal lines have different impedances (different transmitter drivingimpedances or different receiver input impedances), a differential voltagewill be superimposed on the received signal. A similar effect occurs if theactive signal lines have different lengths.

Capacitive coupling from the shield: Although the grounded shield pro-tects the active signal lines from external capacitive coupling, it cannotprotect them from itself. If significant shield currents flow in responseto external sources of interference, these will produce voltage drops alongthe shield which can be capacitively coupled into the internal signal lines.Any differences in the levels of capacitive coupling, or mismatches in theimpedance associated with the two signal lines, can produce interferencecomponents in the differential waveform presented to the receiver. Inview of this, you are still strongly discouraged from connecting the shieldto both the transmitter’s and the receiver’s ground reference point. Asin the single-ended case, it is always best to tie the shield only looselyto ground at the receiving end. At the transmitting end, the shield isconnected directly to ground.

3.3 EMI Suppression Baluns

A useful course of last resort in suppressing EMI on transmission lines is theferrite balun. Baluns are just small inductor/transformer cores, whose originalmain purpose was the construction of impedance matching transforms for analogvideo applications. With the advance of high frequency electronics, however,baluns are seeing greater use in EMI suppression. The idea is to wind somenumber of turns (typically only one) of the transmission cable around the ferritecore. Split balun cores make it possible to do this without actually disconnectingany of the leads.It is important to make sure that all conductors involved in the transmission

line are wound around the balun former together. This means that the EMF’sinduced by magnetic flux in the core should be identical in all signal lines,so that they do not appear as superimposed signal content. These inducedEMF’s appear only when a net current flows in the cable, which is exactly whathappens when currents circulate in the path formed by the ground conductor(or shield) and earth — see Figure 10. As mentioned, the generated EMF doesnot superimpose any net differential signal at the receiver, but the resulting


inductance does serve to raise the impedance of the ground/shield conductor,allowing potential differences between the receiver and the transmitter to bebridged without excessive currents.At very high frequencies, eddy currents in the ferrite core become large,

producing significant power losses. Components which lose power are equivalentto resistors from a circuit model perspective. Resistive losses in an inductorare not normally desirable. For EMI suppression, however, this is actually adesirable phenomenon. Very high frequency components are absorbed in theferrite core, which reduces the amount of EM radiation which can be emittedby the transmission line.

4 Methods to Avoid Being a Source of EMIIn this section, we are concerned with methods for minimizing the amount ofEMI that an electronic product produces. As in Section 2, we can divide thevarious mitigation strategies into primary, secondary and tertiary methods. Pri-mary methods relate to the choice of circuit components and layout; this is thetopic of Section 4.1 below. Secondary methods control the EMI emitted at theinterfaces between sub-systems or products; this is the topic of Section 4.2. Asbefore, tertiary methods involve shielding. Taking the perspective of a victim,Section 2.3 discussed shielding for sensitive sub-systems and products. Takingthe perspective of an EMI source, the same arguments show that shielding canbe used to contain or absorb sources of electromagnetic interference. Due tothe symmetry of the source and victim problems, there is no need to considertertiary mitigation methods again in this section.

4.1 Primary control: components and layout

Many of the methods described in Section 2.1, for mitigating EMI as a victim,also reduce the amount of EMI which a circuit produces. In particular, thefollowing principles should be applied:

Avoid circuit loops with high dI/dt: Current flowing in loops generates amagnetic field. When the current alternates, an alternating magnetic fluxis produced in other parts of the circuit and external electronic productsleading to inductive coupling of EMI. In fact, the alternating magnetic fieldis an origin of propagating EM waves. Therefore, you should endeavourto avoid loops when laying out conductors for signals with strong highfrequency components.

Avoid large conductor surfaces with high dV/dt: Conductors with largesurface areas can be the source of capacitively coupled EMI. Of course,this is relevant only to the extent that the conductor’s electric potentialcontains strong high frequency components. Large surface areas arise inthe context of long conductors, as well as wide conductors.


Avoid sharing conductive paths with sensitive signals: Conductivecoupling through shared power or ground paths has been adequatelydiscussed already in Section 2.1.

In addition to the above-mentioned points, it is worth noting that non-linearities in electronic components present a source of unintended high (orlow) frequency components. As a simple example, consider a sinusoidal volt-age waveform applied to a component (e.g., a resistor) with a square-law non-linearity; this non-linearity produces currents with frequency components attwice the source frequency and at DC. In general, non-linearities allow for inter-modulation of the various frequency components present in your designed prod-uct. These modulation terms may escape the EMI mitigation measures youdesign for the principle frequency components in your circuit.

4.2 Secondary control: interface filtering

In Section 2.2, we noted the importance of filtering signal and power supplyinputs to your product, so as to remove unwanted noise and interference com-ponents which lie outside the frequency range of interest. In some cases, asimple R-C circuit is sufficient, in others power efficient inductor-capacitor filternetworks might be required. From the perspective of an EMI source, filteringis also important.For generated signals, such as those sent over transmission lines, the purpose

of filtering is to minimize unwanted frequency content produced by the outputdriver. When the signals are digital, the chief cause of unwanted high frequencycontent is unnecessarily high slew rates between the low and high logic voltagelevels. Slew rates in digital transmission lines should therefore be carefullycontrolled. A simple R-C filter may suffice. Alternatively, the line driver’scurrent driving (pulling and pushing) capabilities should be matched to thecapacitive load imposed by the transmission line or an explicit load capacitor.If high slew rates are unavoidable over long transmission lines, a differential

line driving strategy can be employed, as shown in Figure 14. In this case, theEMI generated by each of the two signal lines should roughly cancel. Techniquessuch as these are already starting to enter into the design of high speed memorybuses for digital applications.Power supplies in electronic products are also a source of considerable EMI

on the mains power lines. Π filters, such as that shown in Figure 8, serve both tominimize susceptibility to external EMI sources and also to minimize the amountof EMI which escapes from your circuit into the mains. The rectification processitself, however, produces strong current switching transients at 100 Hz (for a50 Hz mains power system). For high power rectifiers, input filtering may berequired to smooth out these switching transients. This generally requires theuse of inductors, as well as capacitors. Figure 15 illustrates one simple powerconditioning filter.


GND

VCC

regu

late

d ou

tput

PowerConditioning Filter

mai

ns p

ower

inpu

t

LM7805in out

ref

GND

VCC

regu

late

d ou

tput

PowerConditioning Filter

mai

ns p

ower

inpu

t

LM7805in out

ref

Figure 15: Mains filtered power regulator to reduce EMI emissions to the mains.

5 Regulatory IssuesAcknowledgement: This final section is taken in large part from:

“Design for Electromagnetic Compatibility,” lecture notes prepared by R.H. Mondel, W. H. Holmes and A. P. Bradley.

A strong regulatory framework exists in all western countries, to control levelsof EMI produced by saleable products (source perspective), as well as the abil-ity of certain types of products to operate within an environment containingprescribed levels of EMI (victim perspective).In Australia, this is administered by the Australian Communications Au-

thority (ACA), which was formed on 1 July 1997 as a merger between AUSTEL(Australian Telecommunications Authority), which was the telecommunicationsindustry regulator, and SMA (Spectrum Management Agency), which was re-sponsible for radio frequency spectrum management and radio communicationslicensing.In the USA, the relevant standards are administered by the FCC (Federal

Communications Commission), while in Europe they are administered mainlyby the IEC (International Electrotechnical Commission).In Australia, complying equipment receives a C-tick. The equivalent in Eu-

rope is the CE mark. There are severe penalties for placing non-complyingequipment on the market in all countries.Most of the standards attempt to control electromagnetic emissions. There

are also some standards (especially in Europe) which specify levels of immunityof equipment to incidental EMI from other sources. These are especially rel-evant if the equipment is safety critical (e.g., medical electronics and aviationelectronics). There are specialized firms which can carry out tests of compliance.The main Australian standards are summarized in Table 1.


Table 1: Regulatory standards governing electromagnetic compatibility in Aus-tralia.Standard EMC/EMI Products CoveredAS1044:1992 Emission Household electrical appliances, portable

power tools and similarAS2064,1&2:1992 Emission Industrial, scientific and medical radio

frequency equipmentAS3548:1992 Emission Information technology equipmentAS4052:1992 Emission Microwave ovens for frequencies above 1

GHzAS1053:1992 Emission Sound and television broadcast receivers

and associated equipmentAS2557:1992 Emission Vehicles, motor boats and spark ignited

engine driven motorsAS4051:1994 Emission Electrical lighting and similar productsAS4251:1994 Emission Generic emission standardsAS4053:1992 Immunity Sound and television broadcast receivers

and associated equipmentAS4252:1992 Immunity Generic immunity standard

chapter8 emc

Education

sources of emi

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electronic product tooperate

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