5

PHASE

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Phase Noisement Seminar

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WELCOME TO THEHEwLErr ftit PAcKARD

RF and ltlcrowavePhase Nolee lleasurement Semlnar

It,rl \ lt \so(f)l \I \---

2MH?

Welcome to the Phase Noise Measurement Seminar.Tbday, measuring and specifying phase noise has becomeincreasingly important as phase noise is often thelimiting factor in many RF and microwave systems. Bothoscillators and devices have phase noise associated withthem that must be measured.

PHASE NOISE CHARACTERIZATIONOF SOURCES AND DEVICES

Typicol Sources: Crystol Oscil lotorsYIG OscillotorsSAW OscillotorsDROsSynthesizersCovity Oscillotors

Typicol Devices: AmplifiersMult ipl iersDividers

WHAT ELSE?

We want to encourage interactive discussionthroughout today's seminar so that we can all share yourmeasurement problems and experiences. We hope tobenefit, too, with new applications awateness,measurement conditions, and measurement technique. Tbstart, we'd like a list from you ofissues and concerns thatyou'd like to see resolved in this seminar and in particularthe types ofsources or devices that you must characterizeand their frequency ranges.

/

SEMINAR AGENDAL Bosis of Phose Noise

$ Wf'y is Phose Noise importont?Whot is Phose Noise?Whot couses Phose Noise?Quont i fy ing Phose Noise

l l . Meosurement Technioueson Sources

1. Di rect Spectrum Method2. Heterodyne/ Counter Method3. Phose Detector Method4. Freguency Discriminotor

Method5. Summory of Source

Meosurement Techniques

l l l . Meosurement of Two-PortPhose Noise (Devices)

lV. Phose Noise Meosurementon Pulsed Corriers

V. AM Noise Meosurement

IMPORTANCE OF PHASE NOISEVARIES WITH APPLICATION

These are the topics that we'll discuss today. Wherepractical, we shall demonstrate the measurementconcepts using both manual and automatic systems. Weshall also try to look more at the practical aspects ofthesemeasurements; many mathematical derivations are lefbto the expertise in the list ofreferences at the end ofyourhandout. Also note that in the back ofyour handout is aglossary of symbols used in the seminar. (This glossarywill also be useful in your further reading of thereferences, as some authors use different symbols for thesame parameter.)

WhV is Phase Noise Important?

There is obviously a difference in the short-termstabilities of sources. But what is thereason to quantify this difference? Primarily, as we shallsee further, short-term stability is often THE limitingfactor in an application. The three applications shownhere all REQUIRE a certain level of performance in short-term stability. Though the required level ofperformancediffers, short-term stability is crucial in each application.

Short-term stability is not a parameter that comes forfree with good design in other areas. In fact, it is one ofthe most expensive parameters to design for. Becausethese three applications require different levels ofperformance, it is very important to quantify andmeasure the required short-term stability, and to thenchoose the right source for the application.

-

LOCAL OSCILTATOR PHASE NOISEAFFECTS RECEIVER SETECTIVITY

IN A MULTI-SIGNAL ENVIRONMENT

WantedSignal

Receiver lF Bandwidth

A high performance superheterodyne receiver serves asa good example for illustration. Suppose two signals arepresent at the input ofreceiver. These signals are to bedown-converted to an IF where filters can separate thedesired signal for processing. Ifthe larger signal isdesired, there should be no difficulty in recovering it. Aproblem may arise, however, if the desired signal is thesmaller of the two. The phase noise of the LO is translateddirectly to the mixer products. The translated noise in themixer may completely mask the smaller signal. Eventhough the receiver's IF frlters may be sufficient toremove the larger signal, the smaller signal is no longerrecoverable due to the LO phase noise. A noisy LO candegrade a receiver's dynamic range as well as selectivity.

Doppler radars determine the velocity ofa target bymeasuring the small shifts in frequency that the returnechoes have undergone. Unfortunately, the return signalincludes much more than just the target echo. In the caseofan airborne radar, the return echo also includes a large"clutter" sigrral which is basically the unavoidablefrequency-shifted echo from the ground. The ratio of mainbeam clutter to target signal may be as high as 80 dB,which makes it difficult to separate the target signal fromthe main beam clutter. The problem is greatly aggravatedwhen the received spectrum has frequency instabilitiescaused by phase noise either on the transmitter oscillatoror the receiver LO. Such phase noise on the clutter signalcan partially or totally mask the target signal, dependingon its level and frequency separation from the carrier.

CARRIER PHASE NOISEAFFECTS SENSITIVITY

OF A DOPPLER RADAR SYSTE]UI

l'* L--ls-Jl m i t l e r I m \

i6G-",i".".vo, | #-

Simpllclty) I._&{r o = r ' *

Signal Frcm

tOCAt OSCILTATOR PHASE NOISEAFFECTS THE BIT ERROR RATE

OF A OPSK SYSTEM

Stote Diogrom

.ooE

c)'cLoC)o

o.oozc)ao_c(L

(Daa

IS ALL NOISE IMPORTANT TO ALt OFTHE PEOPLE ALL OF THE TIME?

1 0 1 0 0 ' l k 1 0 k 1 0 0 k 1 M 1 0 M 1 0 0 MOffset from Corrier (Hz)

In a Quadraphase Phase Shift Keying system, the IQposition of the information signal on the state diagramdepends on the amplitude and phase information afterdemodulation. Amplitude noise affects the distance fromthe origin while phase noise affects the angularpositioning. Close-in phase noise (or phasejitter in thetime domain) on the system local oscillator affects thesystem bit-error rate.

Phase noise is important in these applications, butwhere the noise is important (i.e., at what offset from thecarrier) differs. This graph shows some typical ranges ofoffset frequencies where noise is important for differentapplications. Because the range ofoffset frequencieswhere phase noise is important changes with theapplication, the type ofsource used also changes.

WHAT IS PHASE NOISE?

Gunn Diode Osci l lator

Synthesized Source

What is phase noise? What is a quantity with astatistical randomness?

T\vo very different sources with many differences inperformance (and difference in cost!). One way they differis in their frequency stability. This difference infrequency stability will definitely affect the type ofapplication that they will be used in. What is frequencystability, and how can we describe the difference infrequency stability ofthese two sources?

Frequency stability is generally defined by twoparameters: long-term and short-term stability. It iscommonly said that long-term frequency stabilitydescribes the variation in signal frequency that occursover long time periods, and short-term stability refers tothe variations that occur over time periods ofa fewseconds or less.

However, the dividing line between long-term andshort-term stability is really a function ofapplication. Forexample, in a communications system, all variationswhich are slower than the narrowest carrier or data clocktracking loop would be referred to as long-term, with thedividing line being a fraction ofa second. On the otherhand, a timekeeping system would observe day-to-dayirregularities as short-term, with a dividing linecorresponding to the length of a mission, which might beseveral days.

Long-term stability refers to slow changes in theaverage frequency with time due to secular changes in theresonator. It is usually expressed as a ratio, Af/ffor agiven period of time - hours, days, or even months. Long-term frequency stability is commonly called frequencydrift, and is usually linear or sometimes exponential.

Short-term stability refers to changes in frequencywhich cannot be described as offset (static error) or drift,but are observed as random and/or periodic fluctuationsabout a mean. They are usually described in terms ofvariations about the nominal frequency that occur overtime periods ofa few seconds or less.

This demonstration illustrates the difference in thefrequency spectrum oftwo sources with different short-term stability characteristics.

DEMONSTRATION

HP 8566 A,/8

Stabil ity Comparison

LONG-TERT FREOUENCY STABIUW

(doys, months, yeors)

o Slow chonge in overoge ornominol frequency

SHORT-TERTI FREOUEI{CY STABIUTY

r t

f 6 @

r(seconds)

o Instantaneous frequency variationsaround the nominal frequency

time

TYPES OF NOISE

m 9 . m E 9 1 t f t

f f i r y t t b E t

o Determinist ic (Discrete)

o Con t inuous (Rondom)

Short-term stability is more familiar to most of us inthe frequency domain. Looking at a signal on an idealspectrum analyzer (one with infrnitely sharp filters andno short-term instability of its own), all of the signal'senerg'y dries not occur at a single spectral line, but rathersome of the signal's energ"y occurs at frequencies offsetfrom the nominal frequency.

Using a spectrum analyzer to observe our two examplesources, it's obvious that the sources differ in short-termstability. How can we describe and quantify thisdifference? What units can we use to compare what wecan visually see here? And once deciding upon units touse, how can we measure these values?

In discussing short-term stability, there are two"classes" offrequency variations - non-random (ordeterministic) and random. The frrst, deterministic (orsystematic, periodic, discrete, secular) are discrete signalswhich appear as distinct components on our idealspectrum analyzer RF sideband spectrum. These signals,commonly called spurious, can be related to knownphenomena in the signal source such as power linefrequency, vibration frequencies, or mixer products.

The second type ofphase instability is random innature, and is commonly called phase noise. The sourcesofrandom noise in an oscillator include thermal noise,shot noise. and flicker noise.

lP ^",

COMPARING SHORT-TERMFREOUENCY STABILITIES

16 dB/

C E N T E R I O . 5 3 IRES Bf

462 g EHzlgO Hz VBr IOO Hz

SPAN 26. S kHzSIP 6 . SO a .c

tt/l h

VilI |l,I 1 . . ,I

[lt,r i il,

trI rl|l 1ll llll ll'r I lil rllltfl!

t l

IDEAL SIGNAL

Yk) = Ao sin Lntot

wha,ro,

Ao ; norninol onpllludo

h = notttinof lro4uonorl

REAL WORLD SIGNAL

r ' t f " rv $) = lr" + e(t) | sin I Ln f^t + 0(t) |

l l v l

whoro

t&) = onplitudo lbrchustions

Q0) = phaso lluotuftions

Before proceeding to the definitions ofphase noise, let'sget a more intuitive feel. If one could design a perfectoscillator, all signals could be described like this. In thefrequency domain, this represents a signal with allenergy at a single spectral line.

But in the real world, there's always a little somethingextra on your signal. Unwanted amplitude and frequencyfluctuations are present on the signal. Note that thefrequency fluctuations are actually an added term to thephase angle term ofthe equation ofa signal. Becausephase and frequency are related, you can speakequivalently about unwanted frequency or phasefluctuations.

Concentrating frrst on the frequency fluctuations, let'ssee what these fluctuations would look like on a signal. Inthe time domain, phase is measured from a zero crossing,as illustrated by plotting the phase angle as the radiusvector rotates at a constant angular rate determined bythe frequency. Random noise processes affect the signalthroughout its period.

Let's look on just one particular time in which the sinewave is perturbed for a short instant by noise. In thisperturbed area, the AV and Lt(or AO) corresponds toanother frequency. These perturbations repeat on eachcycle at a recognizable, somewhat constant repetitionrate. In fact, we will find that there is a signifrcantamount of power in another signal whose period is theperiod ofthe perturbation shown.

Thus, in a sideband spectrum (rms power vs.frequency), we will observe a noticeable amount of powerin the spectrum at the frequency corresponding to thisperturbation, with an amplitude related to thecharacteristics of the perturbations. Thus, frequencyvariations, or phase noise since it is really instantaneousphase fluctuations, occur for a given instant of timewithin the cycle. How much time the signal spends at any ^

f,'n".1ii""-J:x'J,::3i':::H:'.Hil*s,11:ilffle"th"ltfrequency domain.

Phase Jitter

Oscilloscope Display

IN THE TIME DOMAIN . . .

V(l; =4ot,n [2rfo t + Ad(t)]

90"

v0)

270"Angular Frequency

180" 00

IN THE FREOUENCY DOMAIN...

Power Spectrel Denslty

N

Egoq,

fo

ABSOLUTE (TOTAI} NOISE

o Specified on sources or completesystem

Another way to think about phase noise is as acontinuous spectrum of infrnitely close phase modulationsidebands, arising from a composite oflow frequencysignals. A signal's stability can be described as powerspectral density ofphase fluctuations or frequencyfluctuations (and later on we'll see the power spectraldensity of amplitude fluctuations).

Discussions about phase noise can be divided into twotopics: the total or "absolute" noise from an oscillator orsystem that generates a signal and the added or "two-port" noise that is added to a signal as it passes through adevice or system.

Absolute noise measurements on the output signal of asystem would include the noise that occurs when thesignal is generated and the "two-port" noise added by thesystem signal processing devices.

TWO-PORT(RESIDUAI OR ADDITIVEI NOISE

. Specified on devices or subsystem

'TWO-PORT V8. ABSOIUTE NOISE

EXAMPLE: SYNTHESIZER

. Absolutet \ l

"Tho-port" (or residual or additive) noise refers to thenoise of devices (amplifrers, dividers, delay lines). Tko-port noise is the noise contributed by a device, regardlessofthe noise ofthe reference oscillator used. One way tolook at "two-port" noise is how much noise would be addedby a device ifa perfect (noise-less) signal were input to it.The name "two-port" emphasizes the contributed natureofthe noise ofdevices.

A "system" - such as a synthesizer - has both two-portand absolute noise associated with it. The reference signalof the synthesizer, comprising an oscillating element, hasabsolute noise. The synthesizer circuitry - phase lockloops, multipliers, dividers, etc. - have some two-portnoise contribution. The integrated system, thesynthesizer, also has a value for absolute noise, or allnoise present at the output.

10

'TWO-PORT" vs. ABSOLUTE NOISE

I

F_

5o

6z

$d

The absolute noise ofthe reference and the synthesizer,and the two-port noise of the synthesizer are comparedhere. Though the units have not been explained yet, thereare still several important points about this graph. One,typically two-port noise ofdevices is less than theabsolute noise on sources, in particular at higher carrierfrequencies. Second, even with a perfect reference, theabsolute noise ofthe system could never be below its two-port noise level.

What Causes Phase Noise?

THE BASICS OF NOISE GENERATION

Thermol Noise?

Noise Figure?

Phose Noise?

In this section we will briefly look at the basics of noisegeneration. What are thermal noise and noise figure andhow are they related to phase noise?

11

Thermal noise is the mean available noise power perHZ of bandwidth from a resistor at a temperature of TK.As the temperature of the resistor increases, the kineticenerg'y of its electrons increase and more power becomesavailable. Thermal noise is broadband and virtuallv flatwith frequency.

Noise Figure is simply the ratio of the signal-to-noiseratio at the input ofa two-port device to the signal-to-noiseratio at the output, in dB, at a source impedancetemperature of 290K. In other words, noise figure is ameasure ofthe signal degradation as it passes through adevice. What do thermal noise and noise figure have to dowith phase noise?

THERMAL NOISE

w"o, po,nl

s?oolrumAnol!{'z0r0ioploy

truautnqla'f

NP = ttTB

K = Boltznrcn'g congtattT = Ton?sroluro K

B = Bondwidth

ForT = 29oKNp = -zo+ dB!$ftO =

Hz

- t1l dbnHz

NOISE FIGURE

t$/N)in N (s/N)outT

, - (6/N)in I' = @r-r lr, = ,ron

F(dE) = ro bo ($/NJg

I-, qs/N)oullrr, ,sot,

Whot do lhormol no\bo ond nciuo,|igurohouo,, lo do with phoco, no\to?,

The noise power at the output of an amplifier can becalculated if its gain and noise figure are known. Thenoise at the output is given by Nour: FGkTB.

The display shows the rms voltages of a signal andnoise at the output of the amplifrer. lVe want to see howthis noise affects the phase noise of the amplifier.

Using phasor methods, we can calculate the effect of thesuperimposed noise voltages on the carrier signal. We cansee from the phasor diagram that VN,-" produces a LA" ,term. For small AZ.-", LA, .: Vpr-,Ay'so".r.. The totalLA, , can be found by adding the two individual phasecomponents powerwise. Squaring this result and dividingby the bandwidth gives Ss(f), the spectral density ofphasefluctuations, or phase noise. The phase noise is directlyproportional to the thermal noise at the input and thenoise frgure of the ampliflrer.

13

AMPLIFIER OUTPUT NOISE AS AFUNCTION OF THERMAT NOISE

AND NOISE FIGURE

l o - l ^

6P;m1

lo+lm

A0 rm5r

Nout = f cfib

USING PHASORRELATIONSHIPS

XJT

For cmoll Ad;65

L,.rms=#- I rokrl-l ,w

Afr6etotol =Mrff i , /+

ss(r)= ^ry=,f, [*,J

In addition to a thermal noise floor of approximatelyconstant level with frequency, active devices exhibit anoise flicker characteristic which intercepts the thermalnoise floor at an empirically determined frequency f". Foroffset frequencies below f-S6G) increases with f-r.

In an oscillator, the white {o and flicker f-t phasemodulations cause even greater slopes ofnoise spectra.Let's see how that happens. First, add a resonator ofsomequality factor Q to the output of an amplifier. Second,connect the resonator output back to the amplifrer inputin the proper polarity for positive feedback. Third,consider the {o and f-'ofthe amplifrer to be represented bya phase modulator LZwith a perfect amplifrer. Next, anyoscillator will shift frequency in response to a phasechange anywhere in its loop, Lf : LA$12q. Since fe and Qare constants, then phase modulation is converteddirectly to frequency modulation. This makes theirspectral slopes 2 units more negative.

NOISE PROCESSES INAN OSCILLATOR

f o + a f

Af : Ad : e -' 2 0

Slmple Feedback Model

ACTUAL PHASE NOISE

ollcollronurrior,f pl

lo = crrrno,r lruquunoy

o Phose noise "f l icker" oppeors( fc

o Ru le o f thumb: " f l i cker " no ise is* -120 dBc/Hz o l 1 Hz o f fse t

So the oscillating loop itselfwill have noise slopes off '

andf -3. But the buffer amplifier found in most oscillatorsadds its own f and f-'noise slopes to the output signal.

The resulting phase noise plot for an actual oscillator isas shown. The frequency domain response of a sourcewould include terms like Random Walk, Flicker andWhite Phase Noise to describe the slope of spectraldensity for given offsets.

INNOISE PROCESSES

THE FREOUENCY DOMAIN

t -2

R$dom walk Phas (whito FMI

NOISE PROCESSES INAN OSCITLATOR

becomes f -z , f -3

th ruA f - nd ro

A Y 2 0

Slmple Model wlth Bufler Ampllller

15

il.

SEMINAR AGENDA

l . Bosis of Phose NoiseWhy is Phose Noise importont?Whot is Phose Noise?Whot couses Phose Noise?

r) quqnl;lt ing Phose Noise

Meosurement Technioueson Sources

1. Direct Soectrum Method2. Heterodyne/Counter Method3. Phose Detector Method4. Frequency Discriminotor

Method5. Summory of Source

Meosurement Technioues

Meosurement of Two-PortPhose Noise (Devices)

Phose Noise Meosurementon Pulsed Corriers

AM Noise Meosurement

ilt.

lv.

Quantifying Phase Noise

There are many different units used to quantify phasenoise. In this section we will examine the most commonones, how they are derived and how they relate to oneanother.

en4

500{0)

OUANTIFYINGPHASE NOISE

oots\sy(f)

oyctsrt"

Due to random phase fluctuations, in the frequencydomain a signal is no longer a discrete spectral line butspreads out over frequencies both above and below thenominal signal frequency in the form of modulationsidebands. We need a way to quantify this frequencyinstability, or phase noise.

Due to the random nature ofthe instabilities, the phasedeviation is represented by a spectral density distributionplot. The term spectral density describes the energydistribution as a continuous function, expressed in unitsof energy within a given bandwidth. The phasemodulation ofthe carrier is actually equivalent to phasemodulation by a noise source. The short-term instabilitiesare measured as low-level phase modulation of thecarrier. Four units used to quantify the spectral densityare shown.

PHASE FLUCTUATIONSIN THE FREOUENCY DOIIAIN

!\

q,3oo.

to foV(t)=Aosln[2rfs t +A@(l)]

Where 6(t) : rondomphose f luctuotions

PHASE NOISE IN TERMS OFPOWER SPECTRAL DENSITY

^[-..r\_

l

f0 SSg Phow noisoloarr'r;r

36$) Spwtrol donrity ofphoso fluotudtlonc

$61(0 Spoctrol donsity ofI roqnnc,y f luctuotionr

Sy(f) Jpoctrcl &,nifty ol' lroctioncl froquc,noy

fluciuotions

t7

sp(0 oR SPECTRAL DENSITY oFPHASE FLUCTUATIONS

}lfstt lron corrioqt ftzfDemodulote phose moduloted signol

with phose detector

AVout = KCl0in K6='tfrod

0nbpoolrum onolyzor

AVp6s(f) ' X6 l$rrrtfl [vJ

,.,,.-L0'76(i) - Avrrme(fl - svrrro [roo'l- - , - ,' B K6tB K6' LHZJ

$vrr.(O = aehe' powo.r ryectrol dtnsily ol tht'voltoq{, flustuations out of thoohaso' do,'lc,c'lor

s40

/'ra')\;/

A measure of phase instability often used is S6(f), thespectral density ofphase fluctuations on a per-Hertzbasis. If we demodulate the phase modulated signal usinga phase detector we obtain Voot as a function ofthe phasefluctuations of the incoming signal. Measuring V",s on Ispectrum analyzer gives AV"-"(f) proportional b LA, ,(f).Sy*"(f)/K62 gives AZ2,-"(f) which is the spectral densityan equivalent phase modulating source in rad2lHz. Thisspectral density is particularly useful for analysis ofphase noise effects on systems which have phase sensitivecircuits such as digital FM communication links.

Another common term for quantifying short termfrequency instability is Sa{l), the spectral density offrequency fluctuations on a per-Hertz basis. 56(0 can bederived from S6(f) by transforming Af(t) from the timedomain to the frequency domain by_Laplace transform.This gives LK1

" : : f LA(f),-" or Afn $) ;^" : P LAz (f),^,which is the spectral density offrequency fluctuations inHz'lHz. Note So1fl: fzSoff) [Hz"lHz]. Caution must betaken when using S6(f) and Ss(O to compare the phasenoise ofsources at different frequencies,

S6s (0 OR SPECTRAL DENSITY oFFREOUENCY FLUCTUATIONS

sL+6)

lr,7l")

offsot lron oo"ir,nf pt]

5s1O unfu, de,riuod lron s66)

Af(t] = | daf,(t)' ?.n dt

Troncl ormoA into the, troquonoy donoin ....,

, r t f f ) =4unhr)/' ll

Af im6(fl = t'ti'rms6)hzl

sor,(f)= N'rmsff)q

5a1(l) = tt sq$)lHz'lt - ll J l z I

s y (0 oR SPECTRAL DENSTTYOF FRACTIONAL

FREOUENCY FLUCTUATIONS

}ffseft lron corrio,r, I pzf

'yff) isroldlt'dto 560 | dAo(t)Af(t) zr dl

\ ,1 j = T= h

lronslorwd inlo lho lrogvo,ncy domoln .,...

v0=* toct. t t 0

rl i^r$)=#Adi'sr)

s.,6,Ltd'rns(t) , ! ' I r l"yur --io,-l--* sotil

lqJ

sv0

frlLH'J

Sy(f), the spectral density offractional frequencyfluctuations allows direct comparison between sources ofdifferent carrier frequencies. S"(f) is also related to Ss(f)ancl Sa{fl. Using the same Laplace transform approach onAf(tyf" we see that the spectral density offractionalfrequency fluctuations is equal to the spectral density offrequency fluctuation divided by fo2.

SINGTE SIDEBAND PHASE NOISE/ O OR SIDEBAND POWER

WITH RESPECT TO CARRIER LEVEL

tI

{(f) l-\[,r,ll

----.l " l l \

ollott lron corrior, I fF,z)

J (0 - Power DcngiU (Onc Phos6 Modulotlon Sldcbond)

Corricr Powcr H2(l) con b. dcrlvcd from s/ (l) uslng Phoac Modulotlon Thcory

"C (D is an inclirect measure of noise energy easilyrelated to the RF power spectrum observed on a spectrumanalyzer. J (0 is defrned as a the ratio ofthe power in onephase modulation sideband on a per-Hertz basis, to thetotal signal power. J (f) is usually presentedlogarithmically as a plot of phase modulation sidebandsin the frequency domain, expressed in dB relative to thecarrier per Hertz of bandwidth [dBc/Hz]. We will see thatJ (f) can be derived from S6(f) using phase modulationtheory.

19

PHASE MODULATTON AT tm ...

fo - fn fo fo+ fm

Produces sidebonds of fmintervols from corrier.

Amplitude of corrier ond sidebondsore determined by L/ peok,the modulot ion index (P) .

Phase modulation at a rate off* produces carriersidebands spaced symmetrically about the carrier atintervals which are multiples of the modulation rate. Theamplitude of the carrier and sidebands are determined bythe modulation index(p) which is equal to LAp"^x.

Bessel functions relate the carrier amplitude to thesideband amplitude. Here we see Bessel coe{Iicients forthe first 10 sidebands as a function of the modulationindex, LAo"4.. We can relate 56(0 to l(D by making animportant assumption. "C(f) refers to the power densityone phase modulation sideband. We can see from thegraph that the sideband power will be restricted to thefrrst sideband only for modulation indices <<1 radian.For the same restriction, the relative amplitude of thecarrier will always be one, and the slope of the Besselfunction will be 72.

BESSEL FUNCTIONSRELATE CARRIER AMPLITUDE

TO SIDEBAND AMPUTUDEJn(F)

t .0

: 2 n = g

Ope.aling Region

Thus for LA o.^u < < l.we have V""6(fl/V" : YzLA p"uu(fl .Then P""s(f)/P, : %LAi.^u(f). Converting the peak phasedeviation to an rms val-ue and normalizing to alHzbandwidth we have Ln(I) : lzSq(f).

FOR L/pear << I RADTAN

u1':ro', = t, = t, ti,py6) (cinusoidotAg)Vspk 1'

W="*a = l aororortf)v r ' ? t + ' r '

Pssbo =tll7 Mr^r(+)1'?6 + t I

- r aOlr"rtfl lroa;lB -- rrz-

-z T lnr.,

P3e50 |T I B=tHz

Ps59(f)

%

l,aLH'

=dl i l=Ls6t+)

REGION OF VALIDITY OF"

,(f)t(f) : +_

r g t @ l K I B K l 0 0 x r i l f f

L ( f ) ( d B c l H z l v s { t H z l a A i H :

E X T F t 1 O F F . 8 6 { O E X 7 R E F . 3 2 8 K H Z P K B E V F

Caution must be exercised when -C(fl is calculated fromthe spectral density ofthe phase fluctuations because ofthe small angle criterion. This plot of "C (0 resulting fromthe phase noise of a free running VCO illustrates theerroneous results that can occur ifthe instantaneousphase modulation exceeds a small angle. Approaching thecarrier, "C(0 is obviously increasingly in error as itreaches a relative level of + 45 dBclHz at a t Hz offset (45dB more noise power at atHz offset in a 1 Hz bandwidththat the total power in the signal.)

On this graph the 10 dB/decade line is drawn on the plotfor a peak phase deviation of0.2 radians integrated overany one decade of offset frequency. At approximately 0.2radians the power in the higher order sidebands ofthephase modulation is still insignifrcant compared to thepower in the first order sideband which ensures thecalculation of J(0 is still valid. Above the line the plot ofJ (0 is increasingly invalid and Ss(0 must be used torepresent the phase noise ofthe signal.

2L

Oy(?) OR STANDARD DEVtATtoNOF FRACTIONAL

FREOUENCY FTUCTUATTONS(TIME DOMAIN}

oy'(r) = Alon vorionr.o - -t*!

(!r*r-Tr)'

- L fy = outrog|Touer lnloruol i lonq

11 = #of sonplos, 7 = pariod ol ooch sanplo,

. stotistiool nooauru,

r $uondc)

Frequency stability is also defrned in the time domainwith a sample variance known as the Allan variance.o"(t) is the standard deviation offractional frequencyfluctuations Af/fo. A short r will produce short-terminformation, while for a long r, the short terminstabilities will tend to average out and you will be leftwith longer-term information.

There are equations available to translate between thetime and frequency domains. The translations apply tonoise processes having particular slopes, and tend toreverse the independent variable axis. An example ofonesuch equation for a slope of I (f) of f-'is given here.

CONVERSION BETWEEN TIME ANDFREOUENCY DOMAIN

&$) or s6G)

olloot lron corr\or,t pl

l\r lsocondl

For orcnplo,lor slogo ol {tl) asl-L.

ffiv d c r r / | t t

O y ( i ) = ' T T - ' "' t o

. fi acowi so t ronolst obl o

. rnuorling oros

22

Residual FM is a familiar measure of frequencyinstability that is related to Ss(D. Residual FM is thetotal rms frequency deviation with a specified bandwidth.Commonly used bandwidths are 50Hz to 3kHz, 300H2 to3kHz, and 2|Hzto 15kHz. Only the short-term frequencyinstability occuring at rates within the bandwidth isindicated. No information regarding the relativeweighting ofinstability rates is conveyed. The presence oflarge spurious signals at frequencies near the frequencyofthe signal under test can greatly exaggerate themeasured level ofresidual FM since the spurious signalsare detected as FM sidebands.

RELATING RESIDUAL FM TO

6a1ff)

T ."1lHz'lL-r.J

olfsa,t lrorn oorrior,f l*z)

ros 7n = $lal rme frl,qul/nc,y douiotion with'inspr,c,ifiod bsndwidth

Sa1(fl = AlLrncfl = +t s6$)BD

rbru,'Fn = [' ,@

= Io'

,e(0 oR s/(0

23

II. Measurement Tbchniques on Sources

SETII{AR AGENDAl . Bosis of Phose Noise

Why is Phose Noise importont?Whot is Phose Noise?Whot couses Phose Noise?Quontifying Phose Noise

c) t t . Meosurement Techniqueson Sources

1. Direct Spectrum Method2. Heterodyne/Counter Method3. Phose Detector Method4. Frequency Discriminotor

Method5. Summory of Source

Meosurement Techniques

ll l. Meosurement of Two-PortPhose Noise (Devices)

lV. Phose Noise Meosurementon Pulsed Corriers

V. AM Noise Meosuremenr

PHASE NOISE OF TYPICAL SOUFCESFOR MEASUREMENT TECHNIOUE COMPARISON

10 MHzru

ORO AT SMESZEB REE.EUNNINGl G H z 4 1 1 0 f t V m A T O G k

F

Here are four representative sources that will be usedthroughout the seminar in judging and comparing thecapabilites of the four measurement methods.

24

1. Direct Spectrrrm Method

PHASE NOISE MEASUREMENTOF SOURCES

1. Di rect Spectrum Anolys is

Speclrum AnalYzer

O-DeviceUnderTest (DUT)

The simplest, easiest, and perhaps oldest method forphase noise analysis ofsources is the direct spectrumtechnique. Here, the Device Under Tbst (DUT) is inputinto a spectrum analyzer tuned to the carrier frequency,directly measuring the power spectral density of theoscillator in terms of J (fl.

HP makes a number of high-quality spectrum analyzersthat might be appropriate for direct spectrum analysis ofsources. Analyzers covering from sub-Hertz to 22 GHz areavailable, to cover the frequency range of any DUT.DIRECT SPECTRUM

MEASUREMENT CHOICES

20 Hzlo 40 MHz

e-* l-", 3.ssnl synrhes ized1 Hz to 40 MHz

O...-.---.-------.--*l-'tr€o.^l Synrhesized10 MHz to 40 GHz

O-ll-Hp8s6ill100 Hz to 1.5 GHz

G-t-npr56&A/B I synrhesized10 MHz lo22GHz

o-_tl*,6,Al100 Hz to 22 GHz

O---.--.----.------*f l{p 8s66dl synrhesized

25

DIRECT SPECTRU]SMEASURE]UIENT CHOICES

@ffi1lt===il"".IFE:ilrrc

Mffif ,,==r EA ilFF:llIE

o 20 Hz to 40 MHzo 3 Hz RBW min. i@)

o -137 dBm to +30 dBmo Synthesized L0

o 100 Hz to 1 .5 GHzo 10 Hz RBW min. @)o -134 dBm to +30 dBmo Synthesized LO

o 100 Hz to 22 GHzo 10 Hz RBW min. (@)

o -134 dBm to +30 dBmo Synthesized LO

Of all the analyzers listed, the best choices for directspectrum analysis are those listed here. These are allanalog spectrum analyzers with synthesized LO's, andnarrow resolution bandwidths.

As we will see later, spectrum analyzers withsynthesized local oscillators offer the optimumperformance for direct spectrum analysis ofphase noise.

This demonstration illustrates the measurement of thesingle sideband phase noise, f (f), ofa source.

DEI'ONSTRATION

INTERPRETING THE RESULTSP , / f \

t (r) = -j*I! fd3c/srlr s

1. Meosure corr ier level P,

2. Meosure s idebond level P"u6 ( f )

3. Apply Corrections46564 ot 64AMHZ - 01.ec! Spcctrum Mcthod

Ln REF 1O.A den ATTEN 20 dB, _r a d B /

C E N T E R 6 4 6 . 6 4 8 M H zRES B l I kHz VAY 196 Hz

SPAN IOA kHzSIP 3 . OO scc

{l yl,,iltlltu,,

lr!t

W I t , a !

DIRECT SPECTRUM ANATYSISCorrcctlon Factoru

1. Noise bondwidth normolizotion2. Effect of spectrum onolyzer

circuitry

A typical spectrum analyzer display would look likethis. What is "C (0 at a 10 kHz offset for this test source?

Here, the power in the carrier P" is read as + 7 dBm.The marker reads -67 dBm, 10 kHz away from thecarrier. P""6 minus P" is equal to -74 dBc. But is thisequal to J (0?

In the direct spectrum technique, there are twocorrection factors that must be used on our value for P""6minus P" in order to yield "C

(f). First, J (0 is specifred in aone hertz bandwidth, and our measurement bandwidth onthe previous slide was not one hertz. Tlpically, in thedirect spectrum technique, it's impossible (and veryimpractical in terms of time) to make measurements in aone hertz bandwidth. We also must be careful that ourmeasurement circuitry does not effect the quantity we aretrying to measure, and that it measures it accurately.

27

1 .

CORRECTING FOR...

Noise bondwidth normol izot ion

Foctors : 10 log -Pio-B t H t

wnereB,,. ' , is meosurement

bondwidthB 19, is one her tz no ise

bondwidth

Normalizing to aLHz noise bandwidth from a givenmeasurement bandwidth is a simple mattet, involving asimple power relationship. Looking back two slides, whatis our measurement bandwidth?

But J({) requires us to normalize to an equivalent I Hznoise bandwidth. A spectrum analyzer's 3 dB resolutionbandwidth is not its equivalent noise bandwidth. (Noise isdefrned as any signal which has its energy present over afrequency band significantly wider than the spectrumanalyzer resolution BW; ie, any signal where individualspectral components are not resolved.) The noisebandwidth is defined as the bandwidth ofan idealrectangular filter having the sameporuer response astheactual IF filter. How do we find this equivalent noisebandwidth?

BUT-RESOLUTION BAN DWIDTH+ NOISE BANDWIDTH

Analog Spectrum Analyzer

l , . l3 dB Bandwidlh

l-Bn----*l

Bn : No ise Bondwid th

Depending on the desired accuracy, there are threeways to derive the equivalent noise bandwidth. In order ofincreasing accuracy. . .

For a first approximation, most Hewlett-Packardspectrum analyzers have a noise bandwidthapproximately 1.2 times the nominal 3 dB resolutionbandwidth setting. For our example, our correction factorfor a nominal 300 Hz resolution bandwidth setting is25.6 dB.

For a more accurate estimate of the correction factor toequivalent noise bandwidth, you could measure the actualresolution bandwidth ofthe spectrum analyzer, using anaccurate signal source and using the analyzer inzeroscan. (A typical synthesized HP spectrum analyzer has aspecified 3 dB IF bandwidth accuracy of tll%o to -+20Vo.)

Of course, the most accurate way to determineequivalent noise bandwidth is to carefully measure theactual power response ofthe IF bandwidth.

Details on the effect of a spectrum analyzer on noise,including how to measure noise equivalent bandwidth,can be found in AN 150-4. Or, for a simpler solution, thereis available software for some spectrum analyzers toautomatically measure equivalent noise bandwidth.

1 .

2.

1

FINDING EOUIVALENTNOTSE BANDWTDTH (Bn)

First opproximotion...

No ise Bn = 1 .2 x nomino l 83 69

Second opprox imot ion. . .

No ise Bn = 1 .2 x meosured Bs as

Most occurote. . .

Meosure equivolent Bn

FINDING EOUIVALENTNOISE BANDWIDTH

&T"ffiff"*;H:;;

HP 858634

29

CORRECTING FOR...

2. Response of Anolog S/Ato Rondom Noise

Actuol Noise : Meosured Noise + 2.5 dB

In addition to the noise power bandwidth correction,another correction is needed when measuring randomnoise as the spectrum analyzer's detection circuitry iscalibrated for accuracy in measuring discrete signals(sinusoids). In most analog spectrum analyzers, there is alogarithmic IF amplifer followed by a peak detector. Apeak or envelope detector used to measure random noiseresults in a reading lower than the true rms value of theaverage noise (typically about 1.05 dB lower). The logshaping tends to amplify noise peaks less than the rest ofthe noise signal, resulting in a detected signal which issmaller than its true rms value. The correction for the logdisplay mode combined with the detector characteristicsgives a total correction for HP analyzers of 2.5 dB, whichshould be added to any random noise measured in the logdisplay.

TYPICAL MEASUREMENT RESULTSn / r \

!(f) : rs-sb\r) laec/nzlh

! .$ ) : Pn . - (Foc to rB ) + c n - P"

1 0 k H z 1 0 0 k H z

Pm : Meosured sidebond level - dBm -

Bs6s : Resolut ion bondwidth - Hz

Noise power bondwidth

- B n = 8 3 6 s x 1 . 2 =

Foctors = 10 Log (8" ) : d B

2.5 dB 2 .5

- d B m -

_ d B c _Hz

-_Hz

Cn : Rondom signol correction

Ps = Corrier level

! r + : -

That was so easy there's got to be a catch. The directspectrum technique is indeed the simplest ofall phasenoise measurement techniques, but is has severallimitations as listed here.

DIRECT SPECTRUMMEASUREM ENT LII ' ITATIONS

Connot seporote AM ond /M

FM (or lM) AM

AM or QM?

A spectrum analyzer cannot distinguish amplitudemodulation from phase modulation or frequencymodulation. The definition of "C(0 is the power measuredin one phase modulation sideband divided by the totalcarriei power. In order for a valid number for .C (f), thesideband power measured must be phase modulationsidebands only. Ifthe source has high AM noise, thiscondition will not be met. Thus, for accurate phase noisemeasurements using the direct spectrum method, thesource's AM noise must be <<ZM noise.

DIRECT SPECTRUTI MEASURE]UIENTS

AREAS OF CAUTION

1. AM << pl,tl

2. Ronge ond resolution

3. Sweep time

31

The primary limitations to the direct spectrum methodare related to the spectrum analyzer's dynamic range andresolution. From the spectrum analyzer simplifred blockdiagram, what elements of the spectrum analyzer wouldaffect range and resolution?

In measuring random noise, the 3 dB BW and shapefactor can limit the resolution of discrete noise. As anexample, a typical source has line spurious at 60 and 120Hz. If the spectrum analyzer has a minimum resolutionBW of 300 Hz, these discrete signals will not be resolved,but instead will appear incorrectly as a high level ofclose-in noise.

Even with a narrow resolution bandwidth, the shapefactor of the spectrrrm analyzer will limit the resolution ofclose-in noise, or any noise close to a high level spur. Forexample, let's say we chose aI}Hz RES BW, with a shapefactor of 11:1. If the source's phase noise 100 Hz away fromthe carrier was down - 100 dBc (a very reasonablenumber for an RF carrier), the filter width 60 dB down is110 Hz, and the noise will be hidden under the skirt of theresponse to the carrier.

WI{AT DETERTINES RESOLUTION?

lF Fi l ter Bondwidth

lF Filter Type ond Shope

LO Stobil i ty - ResiduolFM

LO Stobil i ty - Noise Sidebonds

RESOTUTION OF DISCRETE NOISEAND CLOSE-IN NOISE

IS AFFECTED BY SHAPE FACTORAND 3dB BANDWIDTH

l+l 60 HzLine Spur

I

l l30 HzBW_

Itl

/ I t;\

"/, - r r l

-10 Hz--

BW

RESOLUTION OF CTOSE-IN NOISEIS AFFECTED BY S'A RESIDUAL FM

l l ftI!

Locked

d l , l r

qqi l , t [ "

Unlocked

The other factor in resolution is related to the frequencystability ofthe analyzer's local oscillator. As a frrstmeasure of frequency stability, we can look at the LO'sresidual FM. It would be inappropriate to allow ananalyzer to have a resolution BW narrow enough toobserve the analyzer's own instability. This means thatthe analyzer's residual FM dictates the minimumresolution bandwidth allowable. (Which, in turn,determines the minimum spacing of equal amplitudesignals.) In general, only spectrum analyzers withsynthesized local oscillators are very useful for directspectrum analysis ofphase noise.

We already saw that residual FM only gives a generalfeel for the level ofa signal's instability. Even afterstabilization techniques ofthe S/A local oscillatorminimizes residual FM, there is still sideband noiseassociated with the LO. The noise characteristic of the LOis transferred to all incoming signals onto the IF. Thephase noise on the local oscillator can mask the noise onthe DUT that we might otherwise be able to see, if weconsidered only the 3 dB bandwidth and the shape factor.This sideband noise gives the better shape factor ofthesquare-topped frlter less actual value in resolving close-inor low level signals.

RESOLUTION OFRANDOM NOISE IS LIMITED BY

S'A tO NOISE SIDEBANDS

S/A Sideband NoiseDUT Sideband Noise

lF Fi l ter

33

DIRECT SPECTRUMMEASUREMENT LIMITATIONS

Connot meosure close-into o drifting source

C F : 1 0 G H ZSpan: 10 kHz

Ewoofilno (10 KHz 5pon, 100 Hz Bg 69) z A boc,

Direct Spectrum Analysis is useful for measuringsources with relatively high phase noise, but the sourcemust be fairly stable (long-term wise). That is, the directspectrum technique cannot measure close-in to a driftingsource. The closer-in you desire to look, the narrower theresolution bandwidth, and the longer the sweeptime. On afree-running source with even fairly good driftcharacteristics, this drift during the sweep on thespectrum analyzer will yield inconect phase noise data.

For example, a good cavity oscillator at 10 GHz willtypically drift 30 kHz/min. Spanning as much as 50 kHz(for an offset frequency >10 kHz from the carrier) with a700 Hz resolution bandwidth will typically take at least20 seconds. By that time, the DUT will have drifted 10 kHz!

Here are some typically obtainable spectrum analyzetnoise floors as a function offrequency, as well as the noiseofsome typical sources. As you can see, the directspectrum technique isn't extremely sensitive, it is mostuseful for measuring stable sources with relatively highphase noise at low offsets.

NOISE FLOORS OF SEVERALHP SPECTRUM ANALYZERS

FF€€-RUNNING

Ylt::"6YNfiESIZER

1lg:DBO AT

_'*.-10 MfkXTAL

t

d3oFE

3p

6z

Eo

DIRECT SPECTRUM METHOD

ADVANTAGES.Very f lex ib leo S imp le to useo Disploys l .(f) directlyoAccurote ly d isp loys d iscrete

s igno ls s imu l toneous ly

DISADVANTAGES

oCon' t meosure sourceswi th h igh AM

o Con' t meosure lownoise sources

o Con' t meosure c lose- inon o drif t ino source

Direct Spectrum Anolys is is opt imumfor meosur ing o source wi thre lot ive ly h igh noise ond low dr i f t .

In summary, the direct spectrum technique formeasuring the phase noise on a carrier is an easy, simpletechnique. It displays "C (0 directly, and also accuratelydisplays the discrete signals simultaneously.Unfortunately, it can't be used to measure very clean(spectrally pure) sources, nor can it measure'noisier'sources that have either high AM or drift. It is however,perfect for measuring the phase noise on a stable(typically synthesized or phase-locked) source withrelatively high noise sidebands.

Ib improve measurement sensitivity an alternate directspectrum technique can be used which consists ofaselective receiver and a low phase noise local oscillator(LO). HP offers a very selective receiver, the HP 89018Modulation Analyzer (with Option 030 High Selectivity)and a number oflow noise signal generators which can beused as the external LO.

DIRECT SPECTRUIf, ilIETHOD

Selective ReceiverHP 89O1B

Selective Receiver

o-DeviceUnder

Control

DUT

G*

A ,,r"rr,,,'",.,

!-o

35

iIEASUREIUIENTS WITH ASELECTIVE RECEIVER

L ( fm)

r, _*l I-*f F- 2.5kHz

The measurement procedure for making phase noisemeasurements with a selective receiver is very simple.You just make selective power measurements at thecarrier frequency and at the desired offset from thecarrier, calculate the difference between the readings andconvert the result to a lHz bandwidth.

The 89018 with the high selectivity option (Option 030)makes selective power measurements from 10 MHz to73OOMHz, over 115 dB dynamic range with over 90 dB ofrejection and -r 0.5 dB accuracy. The analyzer's measure-ment bandwidth is 2.5 kHz.

The 8901B's measurement performance can be easilyextended up to 18 GHz or 26.5 GHz with either the HP117934 Microwave Converter and a microwave localoscillator. or the HP Ll729B Carrier Noise Tbst Set andHP 8662A or HP 8663A Synthesized Signal Generator.

The measurement sensitivity of the 89018 ModulationAnalyzer with an external LO is very good. The noisefloor of the analyzer is less than -150 dBc/Hz. You can seeon the plot that for offsets less than 10 kHz the HP 89018with the HP 86624/86634 provides the lowest phasenoise performance and for offsets greater than 10 kHz theHP 89018 with the HP 86424/8 provides the mostmeasurement sensitivitv.

MEASURE]UIENT SENSITIVIWUSING THE HP 89018

^ 4 0

- -40

E - r o

' -oo

u - 1 0 0

O - , r ^

? - 1 4 0

" - r o o

@ - 1 8 0

M E A S U R E M E N T S E N S l T I V I T Y . 8 9 0 1 B + 1 0

\

0 . I H z I H r 1 0 H ! 1 0 0 H 2 1 k H z 1 0 h H r 1 0 0 l H z I M H z l 0 M H z

- - - H P 8 9 0 1 8 a n d H P 8 s 2 A O F F S E T F R o M C A n R I E R { H z )- HP 89018 and HP 86424/8

SIMPLE MEASUREMENT PROCEDURE

1. Meosure corrier level Ps2. Set o reference2. Meosure sidebond level P5ss4. Disp loy resul ts in dBc or dBc/Hz5. Apply imoge correction

TOTAL TEST TIME < 1O SECONDS

The 89018 modulation analyzer will measure anddisplay the side-band level in seconds. The 8901B'seffective noise bandwidth (34 dB for 2.5 kHz filter) can becorrected for by the instrument, providing direct readingsin dBc/Hz if desired. Because the analyzer does not have atuned front end, a correction must be made for the imagesignal.

When the external LO is set to a given frequency, theanalyzer converts the desired and image signals into theIF passband. The resulting signal in the passband is thesum of these two signals. When making phase noisemeasurements, if the noise at the image frequency (910kHz away) is of comparable amplitude to the noise at theoffset of interest, the image noise will sum into the IF andthe result will be affected. This typically happens at largeoffsets where white thermal noise predominates, and theimage adds 3 dB to the result. Tb calculate the imagecorrection either use the C1 plot or use the C1 equation.

Cr : -10Log.o[10Nt/1o) + lJwhereCy : Image correctionN1 : Ilqagengise_levgl UU

offset noise level

IMAGE CORRECTION

DownConverted

Noise

lmageNoise

455 kHzNarrowBand

Source L.O. FrequencyUnderTest

37

TYPICAL MEASUREMEIIIT RESULTS

! (+\

J l+ \

whereDr M -

n

: :->!q hAc/Hzfr s

: P y * C 1

Meosured s idebondlevel relot ive to thecorr ier converted

, , ,lo d6c/ l1z

dBc/Hz

( the onolyzer wi l ld isp loy th is vo lue)

Ct : lmoge Correct ion-0 (1\ -

lmage Correction1 0Y

I7o

5

4

c l ( d 8 )

1 d B0 . 90 . 80 . 70 . 60 . 5

0 . 4

0 . 32 5

I

+ 1 0 + 8 + 6 + 4 + 2 0 - 2 - 4 - 6 - 8 - 1 0 - 1 2

v T M A G E t N d B

V DES I RED

1. The 8901B Modulation Analyzer measures the totalnoise power in the measurement bandwidth andtherefore cannot distinguish between phase noise andAM noise.

2. Measurements as close as 5 kHz from the carrier canbe made before IF feed through begins to saturate theIF amplifrers and the rms detector.

3. In addition, any discrete signals such as spurioussidebands, may be interpreted incorrectly by thismethod. This is because the frlter noise bandwidthcorrection factor (34 dB for 2.5 kHz frlter) applies onlyto distributed noise throughout the frlter passband.Discrete signals therefore will be measured as 34 dBlower than the actual value.

DIRECT SPECTRUM MEASUREMENTSUSING A SELECTIVE RECEIVER

Areas of Caution

1 . AM << QU

2. Offset

3. Resolut ion

In summary, the direct spectrum technique using aselective receiver is fast and easy to use. There is no needto phase-lock signals. The data is displayed conveniently,either in dBc or dBc in aLHz bandwidth. Noise at a givenoffset is measured and displayed in real-time (measure-ment rate = 5 readings/second). The analyzer onlyrequires an exteral LO and testing can be fully automatedusing an external controller. Although the analyzer cannot separate AM and PHiM noise, total SSB noise,regardless of type, is often what is important whenanalyzing communication systems.

DIRECT SPECTRUM ]UIETHODUSING SELECTIVE RECEIVER

a

a

o

a

o

ADVANTAGES

Fost/EosyLow noise f loorDisploys l . (f) directlyEosily outomotedModerote cost

DISADVANTAGES

Con' t meosure sourceswith high AM

2.5 kHz resolutionOffsets ) 5 kHzCorrect for imoge noise

o

a

o

39

2. Heterodyne/Counter l\frethod

The second technique for making short-term stabilitymeasurements on sources is commonly called the'time-domain' technique (or heterodyne frequency countertechnique). The Device Under Tbst and the Reference aredownconverted with a mixer to a low IF frequency, or beatfrequency, Ub. Then a high resolution cou.tle" repeatedlycounts the IF signal, with the time period between eachmeasurement held constant. This allows severalcalculations ofthe fractional frequency difference, y, overthe time period used. A special variance of thesedifferences, called the Allan variance (after its inventor).can then be calculated. The square root ofthis variance iscalled o(r), where r is the time period used in themeasurement. The whole process is generally repeated forseveral different time periods, or r, and o(r) is plotted vs. ras an indication ofthe signal's short-term frequencystability. Note that o(r) is a statistical measure of manvsamples.

Note that because it is really only the IF frequency thatis effected by the instrumentation, the time-domaintechnique is essentially a very broad-band method, easilyusable to 18 GHz. There are, however, several importantlimitations of this method. One, the sources must beoffsettable, with good resolution to set up a fairly lowfrequency beat note. Second, the sources must be fairlystable. In order for the measured beat note frequency to beindicative of the short-term stability of the DUT, the long-term stability must be small relative to the beat notefrequency. Third, since the mixer measures the combinedshort-term instability of the DUT and the reference, thereference must be lower in noise than the DUT. Fourth,the effective offset frequencies that the time domaintechnique can measure are limited by the dead time andsampling speed of the counter. In commonimplementations, this limits the measurement to offsetfrequencies <10 kHz from the carrier. Finally, the timedomain technique always gives an accurate measure ofo(r). However, for an accurate measure of phase noise, thenoise of the DUT must be falling rapidly as a function ofoffset frequency. We'll look at this limitation more closelyin a moment.

PHASE NOISE MEASUREMENT

2.

SourceUnderTest'

Ref

OF SOURCES

Heterodyne / Covnter method

= Afo

fo

(Yr*r -Yr) '

2

o Stotisticol meosure ofsource instobility

H ETERODYN E'COU NTER lUI ETHODMEASUREiIENT CONSIDERATIONS

, $ouroos nust boollu,t

, $ourcQ, drill << Y6

, Roquiroe lout noiso ro+oro,noo,

.}lf to,ls < l}KHz

. Noico nrust foll ropidly wilh incra,osinjln

The time domain technique is a very sensitivemeasurement method for measuring close-in phase noise.It was the first measurement technique that was usablefor measuring at equivalent offset frequencies ofless than1 Hz from the carrier (it's typically usable to an offsetfrequency of 0.01 Hz!). Because of its inherent reading inthe time domain, this technique is very useful where theapplication problem is best thought of in the time-domain,or where very close-in noise is critical (such asmeasurements on time standards for navigation systemsor for digital communications applications).

For comparison, the sensitivity of the direct spectrumtechnique and the time domain technique are comparedhere. The contribution of the time-domain technique forclose-in measurements is easily seen.

H ETERODYN E'COUNTER II ETHOD

SYSTEM SENSITIVITY

-24

;

3 8 0

4 .^"

140

0 l H z l H z 1 0 0 H 2 l k H z t 0 k H z l @ k H z

0FtStT F80tC ln f r l tR lN. l

1 MHz

COMPARISONOF SYSTEM SENSITIVITY

FREE.RUNNING

Ylti::"SYNTHESIZER

.^]1_':DRO AT

_v_10 MHz

_1',_

tE3o

s

z

?

47

H ETERODYITI E,COUNTER iI ETHODLIMITATIONS FOR MEASURING

SYNTHESIZED SOURCES

5t 7t

Sompl ing the counter in oprescribed monner resultsdigitol f i l tering process.

t n o

REF

Returning to the limitations of the time-domainmethod, remember that it requires sources to be verystable (ie, primary or secondary standards or sourceslocked to standards). However, it also has certainlimitations for measuring synthesizers. This arises fromthe fact that sampling the counter in a prescribed mannerresults in a digital filtering process on the noise oftheDUT. A digital filter, as well as having the filtering effectat f, also has the equivalent filter response at 3f, 5f,7f, . . .

This infrnite digital filter integrates in the noise atot^her offset frequencies. If the noise is falling as {q, fr, orF',the integrated noise from the inflrnite digital filter isnot significant. The total noise reading will be higher, butwithin normal error limits.

However, if the noise is falling only as I-1 or f , thisnoise contribution from the other responses ofthe filterwill indeed contribute significantly to the measuredvalue, resulting in an unacceptable error.

In most designs of synthesizers, the 'synthesizer knee'usually occurs at offsets >10 kHz from the carrier, theuseable range of the time domain technique. However,care should be taken ifthe bandwidths ofthe synthesizedDUT result in f-l or fo noise components at offsets <10 kHz.

AN INFINITE DIGITAL FILTERINTEGRATES IN NOISEAT OTHER OFFSETS

offset f (Hz )

offset from "o..,""

f (Hz)

For non-offsettable sources (such as primary frequencystandards), there is also a dual-mixer time differenceconfiguration for the time-domain method. Here, acommon difference oscillator produces two beat signals,which are statistically sampled and compared. In thisconfiguration, the reference must still have equal or lowernoise than the DUT. However, due to the common natureofthe difference oscillator, most ofits noise is correlatedout, except for any P noise component.

In summary, the time domain technique is a verysensitive measurement method for close-to-the-carrieranalysis of signals from 5 MHz to 18 GHz. Itsdisadvantages have made it useful primarily formeasuring frequency standards.

HETERODYNE'COUNTER METHOD

ADVANTAGES

. wide input f requency rongeo good sensit ivity close-to-

corrier

DISADVANTAGES

o requires two sourceso sources must be offset. con't meosure of offsets

>10 kHzo l imitotions for meosuring

synthesized sources

H ETERODYN E'COUNTER ]T ETHOD

DUAL MIXER TIME DIFFERENCE(FOR NON-OFFSETTABLE SOURCES)

fo +uoDifferenceOsc.

REF .

o Reference must hove equolor lower noise thon DUT

o Difference osci l lotor must hovelower fo noise

43

Notes

o

o

3. Phase Detector Metlrod

The third method of phase noise measurement iscommonly called the phase detector method (also calledthe two-source technique or the quadrature technique).This technique is becoming increasingly popular, as wewill see that it is over-all the most sensitive andbroadband technique.

PHASE NOISE MEASUREMENTOF SOURCES

3. Phose Detector Method

(Two- Source Technique)(Quodroture Technique)

aa=o@svnn6(f)

K6'

svrms(f) ' pworspc*rc| Attcfy ol

t oll qc I lucllrrlion s (Y' f k)

K6 = lhnt, doltclor constont (rod/v)

rcl'hz

Reference

PHASE DETECTOR METHODTyplcal System Sensltlvlty

FREE-RUNNING

":?1:::"SYN'IHESIZER

1-'511OffO AT'.-:.-

10 MHz

_I1_

iloE:wtl&tcta &En6 Etrm1ail6 epw aSorE oFFSET FROM CARFTERredta€9 A sEw 60tlw * calBnASE 94sE ilwE

I

d

trt

6

2t@

a

The sensitivity of the phase detector method is shownhere. The phase detector method provides good sensitivityover the entire offset frequency range; it can be used tomeasure high quality standards (good close-in noise) orstate-of-the-art free-running oscillators (low broadbandnoise). The sensitivity shown can be obtained with high-level mixers; note that typically using a lower frequencymixer yields higher sensitivity. We will see later thatanother component in the system is often the real limitingfactor in a real-life application.

45

BASIC PHASE DETECTOR METHOD

We can now compare the sensitivity of three of ourmeasurement methods. Notice that the phase detector hasthe overall lowest noise floor. For this reason. with newimplementations, it is enjoying increased use.

The basic phase detector method is shown here. At theheart ofthis method is the phase detector. T\vo sources. atthe samefrequency and in phase quadrature, are inpul toa double-balanced mixer used as a phase detector. TLemixer sum frequency (2fo) is filtered off, and thedifferenc: &euencl is 0 Hz with an average voltageoutput of0V. Riding on this dc signal are ac voltagefluctuations proportional to the combined phase noise ofthe two sources. Thebaseband signal is oft,en amplified,and input to a baseband spectrum analyzer. In piactice, ascope or dc voltmeter is used as a quadrature monitor.

In the phase detector method, the mixer selection isimportant to the overall system performance. The mixermust cover the frequency range of the DUT, and the IFport must be dc coupled and should have a flat frequencyresponse. Since the noise floor sensitivity is related to tiiemixer input levels, high level mixers yield betterperformance. However, be careful to match mixer driverequirements to available source power. Before going onwith the method itself, we first need to understand tfieoperation ofa mixer as a phase detector.

COMPARISONOF SYSTEM SENSITIVITY

FREE-RUNNING':: il i: :"SYNTHESIZER

itv:ORO AT

-1'_10 MHz

_T'_

o

d

c

PI

T\vo signals are input to a double balanced mixer. Thephase term, Z(t), represents any phase fluctuation that isnot common to both signals. The IF output of the mixerwill be the sum and difference frequencies; the sumfrequency is frltered off, leaving the difference frequency.(The sum frequency is filtered offto prevent this largesignal from overloading the LNA or baseband analyzer inthe system. The filter also provides a crude terminationfor the mixer so that instead ofjust reflecting the sumfrequency at the mixer output it tends to terminate thesum at the mixer output.)

Let the peak amplitude of V(t) be defrned as V6o".p(peak voltage of the beat signal, equal to Kr,V.). Now, tooperate a mixer as a phase detector, the two input signalsmust be at the same frequency. This yields an averagedifference frequency of 0 Hz, and the mixer outputs asignal proportional to the cosine ofthe phase differencebetween the two signals.

DOUBLE-BALANCED MIXERUSED AS A PHASE DETECTOR

VL cos o1t

Ve cos [opt + {(t)]

Vlr(t) = KrVRtot

KLVR ooc

V(t) = KrVRooc

whoro

[c,,t^ - rrtt

Ical^ +c,o;t

Icor, - r,l;t

+ p(t)]*

+ Qof+ .....

r o&)l

KL = n\No'r oftidonol

A PHASE DETECTORDOUBLE-BALANCED MIXER USED AS

v(r) 0

V0) = K;lxcoc [Ccl*-COpt , l,nl

if (D3= t;t

v(t) = KrVxcos ir,{

= Ybpoolr* [oo]

47

DOUBTE-BALANCED MIXERUSED AS A PHASE DETECTOR

V (t) = Vspeelioos IOAl]

lcl $ 111 = 0,K + t)roor Ao(t)AV(t) = t V5p*11 dn Ad(t)

for Lgps44 <z o.Lradion,cin AC(t) ? Ad(t)

AVlt; = 1V6psa1 A{(t)

Av(t) :644P19K6 = phoco, dclo,clor constonl

(volts/rodion) = \'l 69uK

lf we also force the two input signals to be nominally90' out-of-phase (phase quadrature), the instantaneousvoltage fluctuations out of the mixer (phase detector) willbe directly proportional to the input instantaneous phasedifference. For small phase deviations, we operate themixer only in the linear region around 0 V dc, and theoutput voltage is related to the phase difference betweenthe two input signals by a constant, Ks, called the phasedetector constant, in Volts/radian.

Since one of our assumptions in operating this system isthat we are operating near the 'Zero" crossing ofthephase detector, it is important that the input sources to thephase detector stay in good quadrature. Deviation fromquadrature will result in an error as shown here.

IMPORTANCE OF OUADRATURE

arror (db) = ?"o tog fooc (mognit udo, olphaso deuiotion from qutdroluro)l

or

cllowabla doviotlon lron 0Y6o=

PHASE DETECTOR METHOD-PROCEDURE

1. Se t -Up

2. Colibrote

3. Estoblish Quodrotureond Meosure

4. Corrections

BASIC PHASE DETECTOR METHOD

1. Set -uo

BosebondAnolyzer

The phase detector method measures voltagefluctuations directly proportional to the combined phasefluctuations ofthe two input sources. Tb best understandthe phase detector method, and how system noise floor isachieved, it is useful to go through the actualmeasurement procedure. There are basically four steps.

This is the "basic" set-up with the DUT and the REFsignals equal in frequency ancl 90'out ofphase (inquadrature) at the phase detector.

PHASE DETECTOR]IIETHOD PROCEDURE

2. ColibrotionA. Beot note method

Offset one source to obtoino beot note (lF)

AV:KaAfl

DUT

PHASE DETECTOR METHODBEAT NOTE CALIBRATION

K/ : Vbpeok

After set-up, you must calibrate the phase detector. Acommon method of calibration is called the'beat notemethod'. One of the input sources is offset from thefrequency ofthe other to produce an IF signal. From thisIF signal, we hope to frnd the slope of the sine wave at thezero crossings, which is equal to K6.

Tlpically, the levels to the mixer are adjusted such thatthe IF is a sine wave (usually by adding attenuation tothe low signal path or R port). Ifthe IF is a sine wave,mathematics tells us that the slope of a sine wave at thezero crossings is equal to the peak amplitude. DefiningVbo""k as the peak amplitude of the beat note orcafibration signal, Vup.,r. : Kz.

Another good reason to add attenuation duringcalibration is to avoid overloading the LNA (low noiseamplifier) or the baseband spectrum analyzer duringcalibration. During the actual measurement of the noisesidebands, the LNA is designed to amplify lower levelsignals, not high beat notes. Also, we will see that inpractice, best accuracy is obtained ifthe spectrumanalyzer settings are not changed during calibrate andmeasure. This can be accomplished by appropriate settingof the calibration attenuation.

PHASE DETECTOR METHODBEAT NOTE CALIBRATION

Spectrum Anolyzer

Known:1) Kl : v bpeok; ̂V( f ) : Kdtg

2) meosu red vo lue i s Vn -_ -

Therefore:

x / : { z V b . . "av(f) : , fz vbrms ap(f)

On our baseband spectrum analyzer, we will see thebeat note (and ofcourse harmonically relatedcomponents). A spectrum analyzer will measure V6r-".Therefore, Ks is equal to u/2Y6^". Defrning our noiseoutput voltage spectrum as AV(f), we see that AV(f) isequal to u/2 V6-" times the phase fluctuations.

There are also other calibration techniques for thephase detector method. One removes the limitation ofhaving to have a sinusoidal beat note. In this calibrationmethod, Ks is computed from the amplitudes of all the IFsignals.

The phase detector can also be calibrated with a knownlevel sideband.

OTHER CALIBRATION TECHNIOUES

B,Non - Sinusoidot lu|noloaF = A6:ffi 6l + 6si, 311'1+ Otin 56l+...)Ko = A-0b+6C- . , .

Known $ide,band

51

DUT

Bosebond SpectrumAnolyzer

r*\

PHASE DETECTOR METHOD .

PROCEDURE

3. Estoblish quodrotureond meosure

r Spectrum Anolyzer meosuresAV'r-"

After calibration (finding Kq),the beat note is removed(sources set to identical frequencies) and the phase isadjusted to set the sources in phase quadrature. Now theoutput ofthe phase detector is a baseband signal,measured by ihe spectrum analyzet,measuring AVrms2.How do we relate these voltage fluctuations to phasenoise?

The phase detector method measures phasefluctuations. Therefore, the output ofthe phase detectoris most naturally expressed in Ss(f), which is thefundamental measure of phase noise. We can also expressthe output in terms of J (f).

INTERPRETING THE RESUTTS

KN0WNTO Av(f) = K0L(6) QWo Dotost'or)

@) LY(l)= {V v6rn1st4(l) (Fron calibrolion)

(3)sr(fl = a|'rns(l)

,ot 1ainitionlllz

THTRETORE:i, sp() = L$L rrr(f) =

[Avrms(r) l, (in rr,l/l uor^, I eona*ittt')L J

'qU)=ttzWr\r*

lor LQrlq<< lrad

AtrL d\

{tt)=tnsgil='h-ff

Let's express these equations logarithmically, tocorrespond to the power readings on the spectrumanalyzer. .C (f) is as shown here; 56(0 (in dB) would beP " " u - P u - 3 d 8 .

CORRECTION FACTORS

2.

Col ib ro t ion At tenuc t ion

Bcndwid th Normo l i zo t ion

3. S/A Effects

As well as the 6 dB'correction'to the two measuredreadings ofP""6 and P6, in practice there are othercorrection factors to take into account.

The BW normalization correction (because the noisesidebands are rarely ever measured in a 1 Hz bandwidth)and the correction for the response ofanalog spectrumanalyzers to noise are already familiar. The correction for'calibration attenuation'arises from the need to have asinusoidal IF signal.

Ifattenuation is added in the R path ofthe phasedetector, any amplitude change on the input will betranslated to the output. Thus, Vtp."r measured duringcalibration (Kz) will be low by the amount of the addedattenuation. For example, if V6o""L was measured as - 40dBm, and if 30 dB of attenuatioir had been inserted duringcalibration (but removed during measurement), then theactual level of K6 during a noise measurement would be- 10 dBm.

4rtpaQ=

INTERPRETING THE RESULTS(LOGARITH]UIICALtY)

.toqly+%"1=' L vir'nc j

2,O l}gAVrru - ZO log Vb |.rlc + lO bg l+

{6) = Pssp(dBrn) powor ol no\so spoctrurninlHz Bondwidth

-Pp (dBm) ?owor of solibrotionbool nolo

-6(dg) oonvorsion lor rns voltJ,oof Dsol signol ondforsa6) ) {s)

53

DEMONSTRATION

Phaoe Detector tlethod

o

oTYPICAL MEASUREMENT RESULTS

START 0 HzRES BI I kHz VBI lg Hz

STOP ISS kHzSIP 3S. S ecc

"640 MHz Reor Ponel vs. 640 MHz Front Ponel"

46624 o t 64OMH2 F.Pone l va R.Pohc l PHASE DETECToR I {ETHOOR€F -e9 .3 dah ATTEN lO dB

Phase Detector ilethod

INTERPRETING thE RESULTS

A . t o k H z l o o r H zr < ( h

{ i i t fagol= totog # = 1s lo9 PseS- l0 lo9Psrs

= Pn?owor of moosurod no'iso

- Pp Powor ol octuol col cignol

- OAL.ATTEN. ollonuolod col gowo'r

- 6 d g

- Bondwidth N0RMALIZATI0N

+ g/A coRREcrroN (if oPPlicoblo)

XO=

Putting in all correction factors leads to our finalequation.

In an actual system, the sources do not stay inquadrature by themselves. They are forced to remain in a90 degree phase relationship by the use ofa second orderphase lock loop in a feedback path to one ofthe oscillators.The error voltage out ofthe phase lock loop is applied toone ofthe sources, forcing it to track the other in phase.Note that this requires one ofthe sources to beelectronically tunable.

' ' REAL-LI FE'' T EASUREl[ ENT

t \ It\lBosebondAnolyzer

55

Since the phase lock loop (PLL) affects themeasurement data so significantly, it is important tounderstand these effects. There are two importantparameters of the PLL - Loop Holding Range and LoopBandwidth.

The Loop Holding Range is defined as the tuning rangeof the source used as the loop VCO. In practice, thistranslates to how far the source under test can drift, andthe PLL can still maintain quadrature at the phasedetector. For example, if the loop VCO has only 25 kHz ofelectronic tuning range, it might be very diffrcult to try tolock up and track a 10 GHz microwave GaAs FEToscillator.

The Loop Bandwidth is defrned as the rates ofphasefluctuations that the PLL can track. Outside of the LBW,the phase ofthe reference and the. phase ofthe DUT arenot correlated. In practice, LBW defrnes the offsetfrequencies where the measured phase noise data out ofthe phase detector represents the noise ofthe DUT.

LHR - LOOP HOLDING RANGE

(DEF)

. LHR : Tuning ronge of sourceused os the loop VCO

(EFFECT)

o Amount thot source under test condrift ond stil l mointoin lock ot theohose detector

f c t a f

LBW - LOOP BANDWIDTH

(DED. LBW : The rotes of phose

f luctuot ions thot thel oop con t rock

(EFFECT)o Determines o f fse t f reouenc ies

where correct ion foctoris needed

s/(f)

t (f)

/ \Jr^ ^r r\ J t t > v L , I

LBW

Since the phase lock loop is such an important part ofthe phase detector method, it is useful to understand moreabout it, and in particular, what determines the actualloop bandwidth. This simplified block diagram of a secondorder phase lock loop (second order indicating theintegrator in the feedback path) shows several essentialcomponents. Ks is determined by the mixer and thesignal input levels. Ko is a function ofthe source used asthe loop VCO. F and K,(s) are dependent on the design ofthe phase lock loop; often, if F is fixed, it's value will beincluded in the value for K"(s).

The open loop gain is defined as the products ofthegains around the loop. The loop bandwidth is defrned asthat frequency that makes the open loop gain equal to 1.

WHAT DETERMINESLOOP BANDWIDTH?

Second order phaso lock loop model

Kd

K o

F

Ko(")

: Phose slope V/rod(phose detector goin foctor)

= VCo slope (Hz/V)- Loop goin stoge- Loop omplifier goin

(function of freq.)

DUT

Loop Filter

WHAT DETERMINESLOOP BANDWIDTH?

Got=

LOW =

K6rKs(6)mKs6

fluor= t

at a LBW = K6 FKq (s) Ke

D I

EFFECTS of LBW

l0d

mtv

-u7tTlRl'

Noise plots showing effects of loop bondwidth setting

The effect of the loop bandwidth on the measured datais easily demonstrated. This spectrum analyzet display isfrom 0 to 100 Hz. The upper trace is the detected noisespectrum output when the phase lock loop had aLHzLBW. The lower trace clearly shows the noise suppressionwithin the PLL BW when the DUT was locked to thereference in a 100 Hz LBW.

What loop bandwidth is needed for a given source? Thebandwidth of the phase lock loop must be chosen largeenough to keep the phase detector in its linear range; ie,such that the phase deviations are less than about 0.2radians. Thus, for low noise synthesizers, with low close-in noise, it is possible to lock up a 10 GHz source with lessthan a 1 Hz loop bandwith. If the close-in noise is high, orif there are many close-in spurious such as line-relatedspurs (which can also drive the mixer out of its linearrange), a wider bandwidth loop must be chosen to reducethe close-in noise. The noisier the source, the wider theloop bandwidth needed. (In practice, ifa source has veryhighJevel close-in noise, it will be easier to make themeasurement using a frequency discriminator.)

Not only does the phase lock loop add circuit complexityto the measurement, it also exacts a penalty. Because aphase lock loop forces one source to track the other sourcein phase, it also effectively suppresses the detected phasenoise ofthe test source within the phase lock loopbandwidth. (Remember, the phase detector outputs avoltage proportional to the phase difference ofthe twoinput signals. Ifthe phase ofone source tracks the other,the output delta phase will be less.) Thus, when makingphase noise measurements with the phase detectormethod, uncorrected measurements are valid only outsidethe loop bandwidth.

o

WHAT'S THE PRICE OF THE PLL?

o Addit ionol correct ion foctor neededof offsets < LBW

s/(f)

t ( f )

LBW

Offset, f

DETIONSTRATION

Phaae Detector Method wlth PLL

TYPICAL MEASUREIIENT RESULTSa6564 v6 A662A o t 64OMH2 PHASE OETECTOR i l€THOD r t th PLL

START O HzRES gU 3Sg Hz VBY l0 H2

STOP IOS kHz

SIP l0O 3cc

HP 86564 ot 640 MHz

Phage Dctoctor t|ethod wlth PlJ-

INTERPRETING thg RESULTS

' t P<rr l0kHZ lOOKHz{tt)[dOo] = lo log - = lo lo9 pssb - l0 109 Ps' 1 5

= P n ?owor ol moosurod noico

- Pb powor of ogtuol col cignol

- oAL.ATTEN. alltnuolod cal powor

- 6 d B

- Bondwidth N0RMALIZATI0N

+ SIA cARREcltoN (it opplicoblo)

x6)

WHAT SETS SYSTET NOISE FLOORIN PHASE DETECTOR TIETHOD?

DUT

t--r-1 LPF LNA ll-lffiIl

REF 8*FF>lHffilI l i - l

Nds2Noi$1 Noise 3

l. Ph.s Detator at l0 GHz

If we've rejected the direct spectrum method becausethe noise floor was not low enough, how can we use thesebroadband spectrum analyzers as the baseband analyzerin the phase detector technique and still get a lownoise floor?

For use as the baseband spectrum analyzer, threechoices are very popular, depending on offset range ofinterest, and whether or not one or two basebandanalyzers are used.

i

o

ilw*

PHASE DETECTOR METHODBASEBAND SPECTRUM ANALYZER

cHorcEst 20 mHz lo 25 k{zr 2-chonnel FFT \Fv

r Noise source for loopchorocterizotion

. rms overogingt 1 Hz BW normolizotion

o 2OuHz to 100 kHz a"cd\o Single chonnel FFt w

r Time ond frequency disploysr Noise source for looo

chorocterizotiono rms overoging

o 20 Hz to 40 MHzr Synthesized LO \F 7r Trocking generotor for loop

cho rocterizotionr Digitol ond onolog overogingc 1 Hz BW normolizotion

MFil]; : r r - r r r

' i l i , l

!!!!!q!! !lr!.JHP 356tA

r i o

PHASE DETECTOR METHODBASEBAND SPECTRUM ANALYZER

cHotcEs

a

a

100 Hz t o 1 .5 GHzSynthesized LO e+'

l t t : : : ; ;. . l : : : 1 '

a

a

100 Hz to 22 GHz LlSynthesized LO $t52

HP 8566A

But whot obout the i r no ise f loors?

Also very popular for use as the baseband spectrumanalyzer are these two broadband analog spectrumanalyzers. Because these analyzers are often alreadyneeded in the test station for other measurements, use ofthem also as the baseband analyzer leverages theequipment requirements. But what about the requirednoise performance?

NOISE FLOORS OF SEVERALHP SPECTRUM ANALYZERS

I

d

I

z

a

Here are the noise floors ofthe local oscillators fromseveral ofthe HP spectrum analyzers listed. Ofcourse, ingeneral, the noise ofa higher frequency spectrumanalyzer is higher than that ofa lower frequencyspectrum analyzer.

61

WHY DOESN'T A BROADBANDSPECTRUM ANATYZER'S NOISE FTOOR

LIMIT SENSITIVITY?Phose detection gives odded sensitivity;

1. Removol of corr ier o l lows LNA use

ond moximum S/A dynomic ronge-

2. Tronslotes noise to bosebondwhere SrlA is most sensitive.

t 0 0 H , l k | z l 0 k f z l m k N r

0 f fs t r fn0 i l cAnn l tB lN l

Several factors combine to decrease the effective noisefloor (or increase the effective sensitivity) oftheanalyzers. In general, using HP's implementations of thephase detector method, and any of HP's synthesized LOspectrum analyzers, the noise floor ofthe combinedmeasurement system will be limited by the noise of thereference source, not the noise floor ofthe analyzer used.

The noise floor of the phase detector method is fairlyeasily measured. It is important to minimize any delaydifference between the two paths, so that the noise ofthesource will be well-correlated out. This leaves the noisecontributions of the mixer and LNA to be measured.

MEASUREMENT OFPHASE DETECTOR METHOD

SYSTEM NOISE FLOOR

Equol deloy in both poths meonssignol 's f luctuot ions st i l lcorre loted of the mixer

CAUTIONS:rAM noise of source must be

sufficiently rejectedo Poth deloys must be equol

to orevent decorrelotionof source noise.

62

PHASE DETECTOR METHODTyplcal System Senaltlvlty

FREE-FUNNING

Y1l:9::'SYI{THESIZER

1]y_0n0 At' .n* . -

t0 Mt+

_:1_

NE.

!

6

o

F

oaz

?E

wtffiltEvErwj$,marcwa&tE oFFSEIFROMCARnTERWsAEwilMFWrcIrc:

The sensitivity of the phase detector method is shownhere. The phase detector method provides good sensitivityover the entire offset frequency range; it can be used tomeasure high quality standards (good close-in noise) orstate-of-the-art free-running oscillators (low broadbandnoise). The sensitivity shown can be obtained with high-level mixers; note that typically using a lower frequencymixer yields higher sensitivity.

IMPORTANCE OFREFERENCE SOURCE

a

a

Svrms is rms sum of noise ofboth osci l lotors

Meosurement determines upper l imi t

Reference tuning ronge importont

The most critical component (or at least usually thehardest to obtain) ofthe phase detector method isprobably the reference source. Since the spectrumanalyzet measures the rms sum of the noise of bothoscillators, the most important criterion for choosing areference source is that its phase noise be less than whatis being measured. The measured noise sets an upperlimit; the measured noise will be the maximum noise ofeither source and at any particular offset frequency thenoise ofone ofthe sources will be at least 3 dB lower.

Though the absolute phase noise is the most importantcriteria for selection as a reference for the phase detectormethod, its AM noise, if very high, will also affect theaccuracy of the measurement. Also, in general, it isdesired to have the reference source act as the loop VCO,as the noise of the DUT can change if it is being used in adc FM mode. If the reference is being used as the loopVCO, its electronic tuning range will in many casesdetermine whether or not a measurement can be made ona given device. Many low-noise sources that might besuitable for use as a reference in a phase detectormeasurement either do not have electronic tuning(typically dc FM) or they do not have suffrcient tuningrange.

63

IMPORTANCE of REFERENCE SOURCE

HOW ltlUCH LOWER SHOULD THEREFERENCE NOISE BE?

orror (db) = to tog (r * ontitol ry)

A standard margin of 10 dB is usually sufficient toensure the measurement results are not signifrcantlyaffected. Ifa reference source with low enough phasenoise to measure the full offset range is not available,several alternatives are available. One option is to useseveral reference sources with sufficiently low noise atspeciflrc offset ranges. Another method would be to use areference source comparable to the source under test sothat the measurement results can be attributed equally tothe noise from each source. (The assumption is made thatboth sources have equal noise, and 3 dB is subtracted fromthe measured value.)

If three comparable sources are available (within about3 dB of each other and all tunable), it is possible to makethree pair-wise measurements and separate out the noisefrom each source. Ifone ofthe sources is appreciably lower(approximately 3 to 6 dB lower) than the others, its lowernoise performance will still be indicated, but its actualnoise information cannot be accurately separated outfrom the higher noise ofthe other devices.

THREE-SOURCE COMPARISON

Determines noise ofprovided noise ofcomporoble (3 to

eocn sourcesource is6 dB d i f ference) .

Meosurement Meosurement

64

Choice ofa reference source depends on the frequency ofthe DUT, the required noise performance, and whether ornot the reference needs to operate as the loop VCO. (Sincephase noise is probably the single most expensiveparameter in a source, its useful to determine how good areference is required.)

Here are some of the best HP reference sources. For RFapplications, the HP 8640dB features very goodbroadband performance for measuring free-runningsources. The HP 8662A and 86634 provides the lowestoverall phase noise of any commercially availablebroadband source, for DUTs to 2.56 GHz. For microwaveapplications, the cavity tuned HP 8684A has lowbroadband noise and 10 MHz of dc FM. The HP 8670family (and related HP 8340 family) are synthesizers withgood close-in noise. And the lowest overall noiseperformance at microwave is provided by the HP tl729Bl8662A.

Moving next to available instrumentation, it isconvenient to break up the required components intothree essential pieces; the reference section, the phasedetector/quadrature circuitry, and the baseband analysissection. We will look at available instrumentation for allthree ofthese areas.

+20

0

! -20o

.9 -40

.! -60

6o -80oF

* -100

zo -120

fi -140o

- r60

- 1 8 00.

PHASE NOISE of someHP REFERENCE SOURCES

1 H z l H z t0 Hz 100 Hz 1 kHz 10 kHr 100 kHz

Oflsel F.om Ca.rier [Hzl

DUT

PHASE DETECTOR METHOD -

IMPLEMENTATION

5 MHz to1 8 G H z

LPF

H P 3 0 4 7 4 +

gDET

REI I

F- HP -}|

9oo i

r I Phose Detector/ r I*Ref *.<1- ^ l. ------)<t- Bosebond Anolvsis ---)I I quoororureu t rcu t r ry |

' I

t l

I HP 3582A or 3561A or117298/8662A ----tl<- 8568A,/8 or 8566A,/8

or 3585A

65

PHASE DETECTOR METHOD

SITPLIFIED HP II72OB BLOCK DIAGRA]T

t lt l

f ! - _ - - - - - - - - - - - - - - - - - i I

One HP implementation of the phase detector method isthe HP Il729B Carrier Noise Tbst Set, which embodies allthe little bits and pieces ofthe phase detector method,including the ability to generate a very low noisemicrowave reference signal, to make measurementseasier. Not a single box solution, it is always used withone or more RF sources and a baseband analyzer.

It represents a modifred phase detector method in thatit utilizes a dual-downconversion to baseband. A low noise640MHz reference signal (available from the HP8662A or8663 Option 003) is multiplied to microwave with a steprecovery diode. A single combline is mixed with themicrowave DUT, to an IF in the range of 5 to 1280 MHz.The double balanced mixer phase compares the IF signalwith the front panel output signal of the HP 8662A. Avariable bandwidth phase lock loop is used to generateerror voltages, to electronically tune the HP 8662A andmaintain phase quadrature at the phase detector.

Tb be able to determine the noise floor of the HP11?298/8662A at any frequency, we need to understandthe effect of multiplication on the noise of a signal. Sincephase noise (or frequency noise) can be thought ofascontinuous modulation sidebands, phase noise increaseswhen a signal is multiplied.

EFFECT OF MULTIPLICATION ONTHE NOISE OF A S IGNAL

lron bossol olqobru lor snoll n,

VsrbV3

i t, =- Y, ry z Yz LQrrqg

^r,ff)= lbl' = -6dg + ZotoQ atluK

-- ' ' l vs l

' l rnIl +r= xl,,lhon

xAl poaK{tlS = '6dB + 2oto9 J:

J/r \

\{ z zoloard t l r l

66

There are three noise contributions in the HP 117298 -the phase noise on the 640 MHz signal (available with HP8662A Option 003), the noise on the tunable signal at thephase detector, and the two-port noise ofthe HP 117298.System noise will depend on the frequency of the DUT,and thus how many times the 640 MHz signal must bemultiplied.l

COMPUTING HPSYSTEM

system (OAc) :r,

1O log (N2 x tOro

117298t8662ANOISE

N t -

! , _* l -

! ^ _t u z -

! _ -d r 5 -

t z l s+ 1010 + 1010 )

Hormonic o f 640 MHz re fe rence

Abso lu te SSB Phose No iseof 640 MHz reference (dBc/Hz)

Abso lu te SSB Phose No ise o f the5 to 1280 MHz output (aBc/Hz)

Two-port noise ofH P 1 1 7 2 9 8 ( d B c / H z )

TYPfCAL HP 117298t8662ASYSTEM SENSITIVITY

FfiEE-RUNNING'::1ll: :*SYNTHESIZER

jt1_TDRO AT

-.,*._l0 MHz

_a_

I

g

tr

6z

I

The 640 MHz signal is used for multiplication becauseof its lower broadband noise level. Note that for lowmultiplication faetors, the broadband noise on the 320 to640 MHz signal can add significantly to the total systemnoise. At higher multiplications (carriers greater thanabout 8 GHz), the multiplied noise of the 640MHz signalis dominant. This multiplication scheme results in about30 dB lower noise at 10 kHz than that available withmicrowave sources such as the HP 8670 or HP 8340families. (At 10 kHz from a 10 GHz carrier. the Hp g6z0and HP 8340 have typical noise ofabout - 93 dBc/Hz; theHP 117298.18662A has typical noise of -123 dBclHzatthis offset!)

67

PHASE DETECTOR METHOD-I1[PLEMENTATION

Phose Lock ing thru theHP 86624 Reference Oscil lotor

Thus, the HP 11729818662A provides the lowest noisefloor for test signals to 18 GHz. It also provides the loopVCO needed for the phase detector method. Depending onthe source under test and how wide ofphase lock loopbandwidth will be needed to maintain the phase detectorin its linear range, there are several ways to phase lock.First, for very stable sources, phase lock can beestablishes through the electronic frequency control(EFC) of the HP 86624 internal 10 MHz referenceoscillator, using the HP 777298 + 10 V phase lock loopcontrol voltage output. This yields a loop holding range at10 GHz of 1 kHz, and a maximum loop bandwidth of about1 kHz also.

For stable free-running sources (cavities, frxedfrequency free-running sources), lock via the HP 8662Adc FM with the I 1 V loop control voltage. This yields amaximum LHR of 200 kHz, and a maximum loopbandwidth of about 50 kHz.

Note that since obtainable HP 8662A dc FM deviationis a function offrequency (and can drop as low as 25 kHz),the obtainable LHR and LBW is dependent on thefrequency of the DUT and the resultant IF frequency. If ina situation where the allowable deviation is only 25kHzand lock cannot be established, frrst change the DUTfrequency ifpossible to yield a new IF. Ifthis cannot bedone, go to another method ofphase locking.

PHASE DETECTOR METHOD-IMPLEMENTATION

Freq Controllo dc FM

Phose lock ing thru the HP 86624 dc FM

68

PHASE DETECTOR METHOD -

IMPLEilENTATION

Phase locklng uslng a thlrd eourcoas the loop VGO

llP 117298 g w

PHASE DETECTOR METHOD-IMPLEMENTATION

to S/A

Phose lock ing us ing thefrequency control of the DUT

For very unstable sources, a third RF source can beused as the loop VCO. In particular, the HP 86408 is agood choice because it features wide dc FM deviation (upto 10 MHz).

A fourth choice for phase locking is via the tuning portof the DUT. Since microwave oscillators typically havemuch more tuning range than RF sources, this methodcan be useful for locking very unstable sources. However,since the phase noise ofa source can be affected by tuning,it is usually preferred to make the measurement withoutusing the DUT as the VCO.

In general, ifthe source is unstable to the point wherephase locking via a 3rd RF source or locking via the DUTseems the only way to lock, it might be easier to use thefrequency discriminator method, in particular if veryclose-in measurements are not desired.

Also check measurement set-up; see if line spurs can bereduced with clean power supplies.

69

CoMPARISON OF LOOP VCO'SAND LOOP BANDWIDTH (LBW)

Tunlng Range ! Max LBW

HP 8662AEFC

HP 8662Adc FM

HP 86408dc FM

DUT

row/to7

+25 tot2OO kHz

+5 MHz

) 1 0 M H z

1 kHz

150 kHz

Summarizes the options for phase locking and theirtypical performance.

Since the method of phase locking uses differentsources, the system noise also changes. In general,choosing a loop VCO with more tuning range increasessystem noise. However, since it is usually noisier sourcesthat require more loop bandwidth, the increaseil systemnoise typically does not limit the measurement.

EFFECT o' METHOD oIPHASE LOCKING

ON SYSTEM NOISE FLOOR

o

=c0

o

o

zo

Ldt.lt -6

1k 10kOlfset From Carrier (Hz)

\ . . , / -HP 1172sBi

\ ,/ / 86624 in dc FM

\ra- "" 8684A/8

\\

\\

\\\_ HP'117298/

8652A in EFC

PHASE DETECTOR METHOD-PROCEDURE

1 .

2.

1

Set-Up

Colibrote

Estobl ish Quodrotureond Meosure

Corrections4.

Tb see how to use the HP L1729818662A in the phasedetector method to make a phase noise measurement, let'sreview the procedure.

DEMONSTRATION

Phere Detcctor tcthodUring thc HP 1l?i2gBl00g2[

TYPICAL MEASUREMENT RESULTS

START 0 Hz sToP lao kHzRES Bt I kH: VBI 39 Hz SIP lO.O .ec

HP 86738 o t 1 0 GHz

Phase Detector MethodUelng the HP 11729818062A

€6738 ot lgcHz - l l729E/8662A PHASE BETECToR tETHoo(EFc)REF -8.5 dB$ ATTEN 10 dB

INTERPRETING tho RESULTS

r t P<rr l0lHz lOOfHz

{tf)l d8o | = lo lo9 : = lo lo9 Pssb - l0 lo9 Ps' rS= Pn?owor ol noocurod noioo

- Pb powor of oetuol col cignol

- oAL.ATTEN. olle'nuolod ul gowe'r

- 6 d g

- Bondwidth NORMALIZATION

+ S/A c.otfiEcTtoN (if opPliooblo)

{(t)

BUT WHAT IF I WANT TOTAKE TEASURETENTS AT OFFSETS

INSIDE THE LOOP BANDWIDTH?

Even with all the right components, and careful designand technique, the primary limitation to the phasedetector method seems to be close-in measurement. Thephase detector method has theoretically good close-insensitivity. How can we make use of this?

It is possible to characterize the effect ofthe phase lockloop on noise, and then to add to the measured value theamount ofthe loop noise suppression. A phase lock loopcan be characterized by injecting flat noise into the loop,and then measuring the response ofthe loop.

PHASE LOCK LOOPCHARACTERIZATION

Vto

Vti

Loop TestOutputv,o (s)

Forword Goin 1

.1 l\| - uoL

DUT

Loop TestInput

v11 (s)

1-Open Loop Go in

73

SYSTEM SET-UP FORLOOP CHARACTERIZATION

For loop characterization, the HP 117298 has loop testports. A random noise source or tracking generator (or

variable sine wave) is applied to the loop test input, andthe loop response can be traced out at the loop test output.The HP 3582A's or HP 3561A's random noise sourceworks well for this. Note that the noise source in the HP3582A or 3561A is bandlimited; thus, the HP 3582A canbe used to characterize loops only 25 kHz wide, the HP3561A, loops 100 kHz wide.

Here is a typical phase lock loop frlter transfer function.The display yields two important pieces of information.First, the LBW can be determined, which designates theoffset frequencies for which an uncorrected phase noisemeasurement can be made. In this example,loop bandwidth is about 100 Hz. Any noise measured atoffsets greater than 100 Hz do not need correction. Second,the amount ofnoise suppression as a function ofoffsetfrequency can be measured and then used to correct themeisured value of noise. For example, at a 20 Hz offset,the loop suppresses the noise by about 10 dB. The level ofphase noise measured at 20 Hz would need to be correctedby this 10 dB for a valid measurement.

PHASE LOCK LOOPCHARACTERIZATION

Mr|G& -l I dl, gTAllJg PlGm

6.7a,/l

l 0c

totv

- t4.3SlARtr 0 Hz,q ?!Hz

Typicol

ilr 3l!i J* Srr rE E

Yr -& 6 G/r

loop filter tronsfer function

74

We now have our final equation for obtaining "C(0values from the phase detector method.INTERPRETING THE RESULTS

t( ' f) : P, (t) meosured noise power

- Bondwidth normol izot ion

- Pb meosured co l s ignol

- Col ottn

- 6dB

+ S/A correction( i f oppl icoble)

* Loop noise suppression( i f oppl icoble)

AUTOMATIC PHASE NOISETEASURETENTS

SYSTEM ADVANTAGESo lmproved productivity ond

accurocyI Hord copy outputo Flexibility

Since the HP 777298 andHP 8662A areHP-IBprogrammable, an automatic phase noise measurementsystem is easily set-up with the baseband spectrumanalyzer of choice. As mentioned, a common system is theHP 117298/8662A/8566A and the HP 3561A if offsetsclose-to-the-carrier are desired.

AUTOMATIC PHASE NOISEMEASUREMENT SYSTEM

HP 30474

Automotic Colibrotion ond Meosurementof t ( f ) . s / ( f ) , S l r ( f ) , Sy( f ) ,m(f )

Of fset Ronge: 0.02 Hz to 40 MHzCor r i e r Ronge : 5 MHz to 18 GHz

ond oboveMeosurement Accurocv: * 2 dB

AUTOMATIC PHASE NOISEMEASUREMENT SYSTEI'

HP 30474

Direct spect rum modeNo ise s idebond mode

(zo az to 4o MHz)Phose no i se mode

(s I , lHz to 18 GHz)3 .

A fully configured hardware/software system for phasenoise measurements is found in the HP 3047A PhaseNoise Measurement System. It can be used in threemeasurement modes, but is most powerful in the phasenoise mode.

76

The HP 30474 consists of the HP 3582A and HP 35854Spectrum analyzers, the HP 35601A Spectrum AnalyzerInterface, the HP 98364 Desktop Computer, and systemsoftware and documentation. The HP 35601A contains amicrowave phase detector, an RF phase detector,baseband signal processing circuitry, and a powerful,variable bandwidth phase lock loop.

The HP 30474 system software not only takes themanual steps out of the measurement, it also enhancesthe accuracy and data processing capability ofthemeasurement.

HP 3047ASIMPLIFIED BLOCK DIAGRAM

HP 3582A20 mHzlo 25 kHz

1.2 lo'18 GHz

HP 3585A20 Hz to 40 MHz

L

5 MHz lo1.6 GHz

0,02 Hu to40 MHz

Phase LockLoop ConlrolVol lage Oulpul

/b-\R1/l@P

HP 3047ASYSTEII SOFTWARE

Bosic Automotiono Bondwidth sett ingso Amplitude rongingr Meosurementso Conversionsr Grophics

Additionol SoftworeBenefits

o Enhonced occurocyo Extended meosurement

copobil i tyo Doto processing

77

HP 30474SYSTEII SOFTWARE

POWERFUL SYSTEM SOFTWARERAISES ACCURACY AND VERSATILITY

o Signol poth col ibrotiono Phose detector colibrotiono Mixer dc offset colibrotionr Phose lock loop

chorocterizotiono Active morkero Line drowingo Integroted noiseo Three source comporison

Increased accuracy is obtained with the 3047A'sextensive'self-characterization'. The sigrral path iscalibrated from the phase detector, thru the low passfilters and amplifiers, and including the phase lock loopresponse. The 30474 uses an algorithm to correct forphase lock loop response, allowing accurate measurementsmany decades inside the loop bandwidth.

The simplest use of the HP 3047A software is in thedirect spectrum mode, or the 40 MHz noise sidebandmode. In the noise sideband mode, an internal source isused as the reference, improving the sensitivity over thedirect spectrum capability of the HP 35854. It is useful onsources to 40 MHz.

HP 3047A < 40 MHz NOISESIDEBAND ]UIODE

ftE#N

Y.il::-TsmE$4R

":-amATtft

t rM&ru

F

3

z

+

78

CONFIGURING THE HP 3047A(PHASE DETECTOR METHOD) WITHARBITRARY REFERENCE SOURCE

HP 3047A

The most powerful configuration of the HP 3047A is inthe phase noise mode. The standard HP 3047A does notinclude a reference source. For state-of-the-artmeasurements, a second device under test or user-definedreference source (one of which must be tunable) can beused as a reference in the phase detector method.

For RF measurements, the HP 86624 or HP 8663A isan excellent choice as a reference source, yielding asystem for measurement of sources at frequencies up to2.56GH2.

AUTOMATIC PHASE NOISEMEASUREMENTS ON RF SOURCES

HP 3047At8662A OR HP 8663A

5 MHz to1280 (2s60)MHz

79

DEIIONSTRATION

PLL Conlrol Voltage

l o DC FM InPUI

Phaae Detector Method Utlng the HP

HP 3047A

- to

TYPICAL MEASUREMENT RESULTS

l , ( l ) E d B c / H z ) v s f [ H z J

HP 86564 ot 640 MHz

Phase Dotoctor tethodUslng the HP 3047A

For microwave measurements, a number of Hp sourcescould be used as the reference, depending on the noisefloor desired. This source would b-e contr6lled manuallv.

PHASE DETECTOR METHOD -

IMPLEMENTATIONHP 117404

HP 117298

HP 8662A Opt 003

The HP Tl740|Microwave phase Noise MeasurementSystem is.a complete Automatic System for phase noisecharacterization of sources, b MHz to 18 GHz. It consistsof the HP 7L7298 Carrier Noise Tbst set, the Hp g6624 (orHP 86634) Synthesized signal generator, the Hp B04ZASpectrum Analyzer System, th;Hp ggB6A DesktopComputer with re_quired_operating system and memory(ordered separately), and iystem soflware, documentationand warranty.

The HP lI740A adds the necessary hardware andsoftware integration of the critical low noise microwavereference source into the standard Hp 3042A SpectrumAnalyzer System, with an overall system specificationand warranty. All standard features and softwarecapabilities ofthe HP B}47Adiscussed on earlier pagesare retained.

The-options permit deletion of an instrument alreadyowned or upgrading to a higher performance instrument.

In this configuration, the Hp lf72gB is used toupconvert the HP 86624, and to downconvert themicrowave DUT.The phase detector, baseband signalprocessing and phase lock loop ofthe Hp 3042A, areemp-loyed to complete the phase detector methodrmplementation.

HP gO47A PHASE1.2 GHz <

DETECTOR METHODf" < 18 GHz

1.2 lo1 8 G H z

A.2nd DUT I

Out

2 to 18 GHz

l 0 M H z l o l S G H z

50 MHz lo 18 GHzor 26.5 GHz

dc FM

dc FM

81

HP 117404

SYSTEM ADVANTAGES

r Bui l t - in low noise microwovereference source ond VCO

r Complete system sPecif icotions

o User- f r iendly ,menu driven softwore

'System documentot ionond suppor t

DEMONSTRATION

Phase Detector MethodUelng the HP 117404

TYPICAL MEASUREMENT RESULTS

Phase Detector MethodUolng the HP 117404

Here's a very detailed isolation ofpotential errorsources. Ofcourse, the largest potential error is therelative accuracy ofthe spectrum analyzer, with theaccuracy ofthe calibration step, and the baseband signalprocessing (Iow pass filters, LNA) flatness also vervimportant.

_ Now, depending on whether the errors are simplylinearly summed, or the root sum of the squares ii laken,or maybe some combination of the two, a phase noisemeasurement using the phase detector method can bemade with better than -F 2 dB accuracv.

PHASE DETECTOR METHODCONSIDERATIONS IN SYSTEM ACCURACY

coNTRtBUroR TyptcAL uNcERTAtNrr (tdB)1. Spectrum onollzer occurocy 0,4-1.s

(relotive)

2. Attenuotor occurcrcy of collbrotion

J. Miemotch uncertointy (osaociotedwith setting CW rcferencc lovel

4. Uncertointy in 12.5 dB correction

5. Accurocy of meosured lF noise bondwidth

6. Relotive lF bondwidth goins

7. Ntolyzer frequoncy reaponae

8. Phose detector flotnege

9. Bosebond signol processing flotness

0.4

0 .15

o,20

o.20

0.05

o,25

o.20

0.s0-1.0

PHASE DETECTOR METHOD

CONSIDERATIONS IN SYSTEM ACCURACY(CONTINUED)

ooNTR|BUTOR TyptcALUNCERTA|NTYGdB)10. Rondom crror duc to O.s

rondomnsss of nolga

I 1. Mlxar dc offcct^uodrqturr O.OJmolntcnoncc

12. Nolsc floor contributlon 0.2( t ra = rd i -15d8)

15. Accurocy of loop choroctcrlzofion O.2

LINAR SUMMATIONRSS

MlnlmumLfNFAR SUMMATION r.2aRSS 0.86

OUTSIDE LOOP BANDWIDTHMlnlmum Moxlmum

5.0E .t 6E0.65 1.53

INSIDE LOOP BANDWIDTH(one decode)

lloxlmum.$.881.53

83

PHASE DETECTOR METHODGENERAL SYSTEM PRECAUTIONS

. Non-lineor operotion of the mixerwill result in o colibrotion error

o Soturotion of the omplifier or SrlA incolibrotion or by high spuriour signols(e.g. line spurs)

o Distortion ol RF-signol yields deviotionof K7 lrom U5 Peok

o Supression or peoking of noise neor thephose lock looP bondwidth

. lmpedonce interfoces should remoin unchongedbetween colibrotion ond meosurement

r Injection locking of the DUT con occur

r Deviotion from phose quodroture(results in lower K1)

r Closely spoced spurious con be misinterpretedos Dhose noise with insufficient S,/Aresolution ond ov€roging

r Noise injected by peripherol circuitry(e.9, power supply) con be o dominontcontributor to Phose noise

o Vibrotion con excite significont noise in DUT

PHASE DETECTOR METHOD

ADVANTAGESo Lowest overo l l no ise f loor

o Wide ronge of o f fset f requencies

r Wide input f requency ronge

o Some AM suppress ion

o Eosi ly outomoted

DISADVANTAGESo Requi res two sources

o Complex i ty

o Close- in meosurements requi reoddi t ionol correct ion

o Diff iculty with locking up high drif trote sources

The phase detector, the PLL, the reference source - allare critical components in the phase detector method.Measurement procedure and system set-up can also bevery important. Here is a list of things to watch out for,and possible sources oferror.

In frnal summary, the phase detector method isenjoying increasing popularity because it has the lowestoverall system sensitivity. It is also the most complexmethoil, but with careful system design, it can be a verygeneral-purpose solution.

84

4. trYequency Discriminator Method

The fourth method of phase noise measurement onsources is called the frequency discriminator method (alsosometimes called the one-oscillator method). In thismethod, the frequency fluctuations ofthe source aretranslated to baseband voltage fluctuations which canthen be measured by a baseband analyzer. K6, thefrequency discriminator constant inYlHz, is defrned asthe translation factor between the frequency fluctuationat the input ofthe discriminator and the correspondingvoltage output. The fundamental output ofthe lrequencydiscriminator can be described by 56(0, the spectraldensity of frequency fluctuations.

There are several common implemenations of frequencydiscriminators, each with its own advantages anddisadvantages. For example, a cavity resonator used as afrequency discriminator can yield very high sensitivity,but typically very narrow input bandwidth. We willconcentrate our discussion on the delav line/mixerimplementation.

FREOUENCY DISCRII'INATORS

o Deloy Line/Mlixero RF Bridge/Deloy Lineo Covity Resonotoro Slope Detector ond

Rotio Detectoro Duol Deloy Line

(Cross-spectrum Anolysis)

PHASE NOISEMEASUREI'ENT OF SOURCES

4. Frequency Discriminotormethod

A f r A V

$61(f) :

$yrhg(f) '

M'r^r0 | url 6vrm6(f)t - t -

b l"r, ) r,dL

?otrtl,r s?e,olral donsity ol tltovoltogo f luctuotions (u7 Hz)

K4 = lroguo'ncty disorininolorconslont (UlHz)

85

TYPICAL SENSITIVITIES OFFREQUENCY DISCRIMINATOR METHOO

FffEE-RUNNING

Y:t::.::'SY{THESIZER':_Y

mo AT1 GHz

l0 MHzru

t

A delay line/mixer used as a discriminator has typicalsensitivity as shown; notice that the sensitivity is afunction of the delay time. The frequency discriminatormethod has very good broadband sensitivity; however,because ofthe inherent relationship between frequencyand phase, the sensitivity of the discriminator methoddegrades as llP as the carrier under test is approached.Because this slope follows the noise characteristic offree-running sources, and because this technique does notrequire a second source for downconversion, it is useful formeasuring sources with large, low rate phaseinstabilities. But because ofits high close-in sensitivity, itis not very useful for measuring very stable sources closeto the carrier.

This graph then cornpares all four measurementmethods with the four "standard" sources that have beenplotted on all previous sensitivity graphs. Remember thatthe phase detector method sensitivity is really only atheoretical value given a perfect reference source. In apractical system, the noise floor ofthe reference sets thesystem noise floor.

COMPARISONOF SYSTEM SENSITIVIW

FFEE.RUNN]NG':: il i: :"SYNTHESIZER

.1lr:DRO AT

_.,*'_10 MHzrTAL

I:9aF

ac3o

Ioz

86

DELAY LINE'MIXER USED AS AFREOUENCY DISCRIMINATOR

I

In the'delay line/mixer'implementation of a frequencydiscriminator, the DUT is split into two channels. Onechannel, sometimes called the non-delay or referencechannel, is applied directly to one port ofa double-balancedmixer, which will be operated as a phase detector. Thischannel is also referred to as the local oscillator channelsince it drives the mixer at the prescribed impedance level(the usual LO drive). The other channel is delayedthrough some delay element, and then input to the otherport of the mixer.

Note that the delay line can serye as a frequency tophase transducer. That is, the nominal frequency arrivesat the phase detector at a certain phase. Ifwe change thefrequency passing through the delay line, then theamount of phase shift incurred in the frxed delay time willbe proportional to the frequency. This delay elementuncorrelates the noise ofthe source arriving at the phasedetector in the two paths. The phase detector thenconverts the phase fluctuations into their voltageequivalent for measurement.

MEASURING S 61(0l l + . I d + l V

rd

-Q

af

LV rns

AVnnr

6M$)

= LTloT4

= ziiidA,fi'})

= KoLQrnt

= KOz.xrdA+rns(d)

=%G)0

_ Avi^rill-@le=,r,

87

THEORY OF DEI.AY LINE FREOUENCY DISCRITIINATOR

Oelay lime t

l ( t ) = lo +Al sin 2zl l

V(t) = vo cos (2r lot '+cos 2' l t )

Carrier with sinusoidal FM lo - Caar'et lrequency

f = F M r c l e

Al - FM devlation

Phase Dit lerence on Phase Detector Ad(l)

ad(t) = 2nto ( t - r) . {cos

2nf(r - ' ) - 2rtot - A} cos 2.t l

L6(tl'-2ntor' f(cos 2.f 0 - r) - cos 2.fi;l

a6(tt = - 2rto' - z f sin "r' [.i"p"r fr -]ll

Phase Detector Oulpul: V(t) = K6 Ad(t)

lor 2nlst=(2k - 1 )f(Phase Ouadrature):

v(r) = K4 2f sin zt' sin 2rt (l -t)

av = K6 2f sin rtr = K42n ttfiJb

lorl<Lu,: s!+"t =1

LV = K62rr L, l

7 I # ] ' Frequencyoiscr iminatorconstanl =K62rr

A more rigorous derivation of a delay line/mixer used asa frequency discriminator is shown here. The importantequation is the frnal magnitude of the transfer response.The sinusoidal output term ofthe mixer responds as sinrfrlnft. This means the output response is periodic inlo-_ 2rrf, and will have peaks and nulls, with the first nullatll' 6. Tb avoid having to compensate for the sin x/xresponse, measurements are typically made atmodulation frequencies well away from the null,Yz or less.It is possible to measure to offset frequencies out to andbeyond the nuII by scaling the measurement results usingthis function, but the sensitivity ofthe system gets poornear the nulls.

Also note that the frnal transfer function (assumingmeasurements made <Yztd) is independent of carrierfrequency. Thus, a 50 ns line at 10 GHz has the samesensitivity as a 50 ns line at 500 MHz.

aTb see the effect ofthe delay element on input

frequencies, note that ifthe differential delay between thetwo channels is zero, there is no phase difference at thedetector output as a swept CW signal is applied to thesystem. Here, the detected output interference disptay isshown when a swept frequency CW signal is applied to asystem which has a differential delay of500 ns. (Thesigrral amplitudes are here assumed to be almost equal')Since there is a two-channel note, there is a null in theresponse every 360 degreess llraHz as shown.

Lastly, the conversion to "C (0 is shown again here.Because the frequency discriminator outputs Soff)directly, when the sensitivity of the frequencydiscriminator system is plotted as a function of phasevariations J(f) or S^(f), the offset frequency squaredterm, f2, in the deno-minator indicates that systemsensitivity will increase by 20 dB per decade as theoffset frequency of the measurement decreases. Thesensitivity gets better until it equals the sensitivity ofthe phase detector at an offset frequency ofl/(2nt)

To vs. USABLE OFFSET FREOUENCY

7d:500 ns

0 t o 4 M H z

fml : 2 M H z l : 4 M H z

_ 1Td

Swept Colibrotion Disploy

" . ( f )

. t ( f )2 f2

88

FREOUENCY DISCRIMINATORMETHOD PROCEDURE

1. Measure Power Levels2. Calibrate3. Establish Quadrature and Measure4. Corrections

Tb better understand the frequency discriminatormethod, let's go through a typical measurement procedureon our test source. There are basically four steps in thedelay line/mixer implementation: 1) set-up and -easrr"epower levels, 2) calibrate, 3) establish quadrature andmeasure, and 4) corrections.

One calibration (ie, establish the transfer coefficientfrom frequency fluctuations to voltage fluctuations)method is to determine Ka from ta and Ks. ra is a functionofthe length and type ofdelay element uied; 16 isapproximately 1.5 ns/foot for cables with polyethelenedielectric. Kq, or the phase detector constant, can bedetermined by establishing a sine wave out of the mixer.The slope of the sine waveat the zero crossings (K6) isequal to the peak amplitude of the signal (forlinear mixeroperation). This calibration method requires an accuratemeasure of the delay, 16. This can be determined bymeasuring the carrier frequency difference, A fn betweentwo consecutive zero crossings on the quadratuie monitor(Afo : I ).

2ra

2. CALIBRATTON METHOD #1

noosuro i4 ond K6

nt /+\ . Avims(f) l rr,15 1 ; ( f l = a t r n 6 t ' / -

L -(K62nrl)' Kd' L itrj

7

id is o funstion of do,lay o,lont'nl uso,d

Td 7 l,Snblfoot for coblos withp oly o,lh o,l on o di el o ct ri c

Kd = YbpooK for sinusoids

89

2. CALIBRATION METHOD #2Find composi te K6 by evoluot ing responseof system to o well-chorocterized input.

t d

SpectrumAnalyzer

fmAfrms Hz

6v

lo tm

I

or

^l l,d

U s i n g :o K n o w no K n o w nr Known

t\d -

s idebond/s ignolBo rdeviotion

A more common calibration method is to determine thecomposite Ka by evaluating the response of the system(delay line, mixer, low pass frlter and LNA) to a knowninput. This can be done by setting up a single FM tonewilh known sideband/carrier ratio, or by setting up FMsuch that the carrier is nulled, yielding a knownmodulation index, or setting up a known deviation (withthe system in phase quadrature). The response ofthesystem to this known value can then be used as areference level.

After calibration, the input signals to the phasedetector are re-adjusted for the phase quadratureconclition (90 clegrees out of phase) if necessary. There areseveral ways to establish quadrature. First, you can varythe frequency of the DUT slightly (perhaps a couple ofMHz is sufficient), until the average voltage out of thephase detector is 0V dc (measured on a quadraturemonitor such as an oscilloscope or DVM). If the DUTfrequency cannot be changed, a line stretcher can beadded to the fixed delay path and adjusted untilquadrature is established. Tlpically, no more than a fewinches ofvariable length are needed. Ifthis is awkward,manual or digital phase shifters can also be used.

Once quadrature is established, the spectral density ofthe voltage fluctuations can be measured on the basebandar'alyzer. Tlpically, the delay line is broadband enoughsuch that even ifthe source drifts, the two inputs to thephase detector will remain in the quadrature condition forthe duration of the measurement. However, quadratureshould be monitored during the measurement, in case theDUT frequency changes significantly.

3. ESTABTISH OUADRATUREAND IUIEASURE

O-D

A.

ii:ii#Estobl ish Quodroture by:1. ) Using phose sh i f ter2. ) Using l ine s t retcher3.) Vorying DUT frequency

Meosure Sv. . r ( f ) on spectrumonotyzer

B .

90

4. CONVERT TO PROPER UNITSAND CORRECTIONS

rPta 4/

sTml ! &E E i l t & V d - k

F r c a &

Av'rms ii I Hz Bcndwidth: 6id.bnnd teutl - Avims- norn|olizdtion to lHz _- g/A corrcction

s^r(n.fu) a,.-. H L

f , i

0

After measuring the spectral density of the voltagefluctuations, the measured values are converted to theproper units and corrected. First, since Sa(fl is defined ona per Hertz basis, the measured noise is normalized to a 1Hz noise bandwidth" The 2.5 dB spectrum analyzercorrection is taken into account if necessary. But how dowe convert the measured noise (in dBm) into units ofHz2lHz?

Ifunits ofSo(fl are desired, the data manipulations areshown here. They look more complicated then they are,because the data has to be translated from log form toabsolute terms and back again several times.

COMPUTING Sar (0 FROMS'A DISPLAY

l, lstoblish Known col, sidr,banAfurr'nr . otoal = - dbc

z.rrcnstnla to $ = onlilog lcot/to = - Hii

A. Find L1661 in rnsHz

A{col ' fm ,T

= - rnsHz

l,lronslolo A{s61 to dDHz

Afa61 (dBHz) = zotog Afcol = - dBHz

5. lltosuro Pool = - d1m

6, find scolo laclor

$F(db)= Llsal (d0Hz) - Ps6g (dgnr) = - db{zld}n

1. P\twuru, P*siss (n l Hz gondwidth)

8. $61(i (dB)' Pn6i5a + SF

9.sn+o @' onr , ,o . . so+r f l (dB) =-E\Hz / l0 HL

91

CONVERT fO T(O IF DESIRED

S"+6)el4\ = :'-dr'/ zll DkHz foo kHz

t0 = nouuredpilsa,lsysl - (dbnr)- delocttd (dBm)

calibrotion lavol'sot sidobondto (d0c)

c,orrio,r rotio

Bondwidth Normclizotion - Ug)s/A coRREcrlo[ (it opptiwbto) 0B)

- ?otoq l^f tu l gg)

/ (l) = - 6Bc/Hz) -

DEMONSTRATION

HP 85668Speclrum AnalYzer

Frequency Dltcrlmlnator Method

Since we are using a spectrum analyzer (whichmeasures power) as our baseband analyzer, it is mucheasier to translate the measured voltages into Ss(f) or "C(0,as shown here. Note the correction factor to translatethe calibration term from the calibration offset frequencyto a given measurement frequency. This term comes fromthe 1/{2 in the translation of So{fl to "C(f}.

TYPICAL IIEASUREIIENT RESULTS45554 o t 646 lHz - OELAY L INE OISCRIMINATOR METHOO

SfARI 6 Hz STOP l9A kH2

RES Sf 1 kHz VBf 30 Ht S IP lO 'O 6cc

HP 86564 ot 640 MHz

Delay UnelMlxerFrequency Dlscrlmlnator Method

Aa REh -39 .9 dBh ATTEN lS dB, -

92

CONSIDERATIONS INSYSTEM ACCURACY

CONTRIEUTOR TYPICAL UI'ICERTAINTY (+dB)

1. Accurocy of powcr lcv€l 0.05meogurementg (relotlve)

2. Spectrum onollzer occurocy (relotivc) 0.4-1.5

3. Attenuotor occurocy of collbrotion 0.06

4. Migmotch uncertointy osgociotcd wilh 0.15setting O{l rcfcrcncc lcvcl

5. Unccrtclnty ln +2.5 dB correction O,2O

6. Accurocy of meosurcd lF nolge bondwldth 0.20

7. Relstlve lF bondwldth goins 0,05

6. Anotper frequcncy r€epone€ 0.25

Here's a fairly detailed analysis ofpotential errorsources, with an estimation of typical values. The majorcontributor to system accuracy is the accuracy ofthebaseband analyzer; this is kept small by using thear.alyzer in a relative mode (ie, by measuring the noiserelative to the calibration level). The flatness ofthebaseband signal processing section can also contributesignificantly to overall system accuracy. Better accuracycan be achieved ifthis contribution is measured and thencompensated for.

Tbking all these inaccuracies into account, an overallsystem accuracy can be determined.

CONSIDERATIONS INSYSTEM ACCURACY

(Continued)

TYPICAL UNCERTAINTYcoNTRTBUTOR (tdB)

9. Frcqucncy dlacdmlnotor flotn $ O.30

10. Borbond algnol proccolng flotncrs 0.50-1.0

11. Rondom crror duc to rqndomnco 0,50of nobc

12. Mlx.r dc offect/quodroturc molr{cnoncc

13. Accurocy of rctllng mod lndaxor rldcbond coriqr

0.03

0.10

TOTAL UNCERTATNTY (tdB)

Lineor SummotionRSS

Minimum Moximum

2,7e z].390.64 1.53

93

USING THE HP 8901AMODUTATION ANATYZER

AS A FREOUENCY DISCRIMINATOR

150 kHzto 1.3 GHz

100 Hzlo 1.5 GHz

100 Hz to22 GHz

HP8660CHP8662/3AHP8672AHP86738./CtDHP 8340AHP 8683A/B/DHP 8684 A/B/DHP 117298/8662A

21.4 MHzlF Out

HP 3582AHP 3585AHP 3561AHP 3047AHP 8568A/8HP 8566A/8

BasebandSpeclrumAnalyzer

21.4 MHz lF Out

10 MHzto 26 GHz

DUT

The simplest frequency discriminator implementationis the HP 8901A. It can be used directly with a basebandspectrum analyzer for sources to 1.3 GHz, or higherfrequency sources can be first downconverted and theninput. Alternatively, the IF output from the HP 8568A/8or HP 8566A/8 can be used to drive the HP 8901A.

The HP 8901A, combined with a downconverter ifnecessary, is a simple solution providing sufficientsensitivitv for manv sources.

TYPICAL SENSITIVIW OF THE HP 89OAUSED AS A FREOUENCY DISCRIMINATOR

FREE-RUNNING

"T1l::*'SYMTHESIZEF

.11:_YD8O AT

-TT_10 Mfh

_1'_

tt'I

E(3

e+E

94

FREOUENCY DISCRIMINATORMETHOD - IMPLEMENTATION

HP 3047A

o d isc r im ino te o f cor r ie r f requency

5 MHZTO

18 GHZ

USER-SUPPLIED

For better sensitivity, the HP 3047A can be used in afrequency discriminator mode (delay line/mixer used as afrequency discriminator). The DUT is externally split, oneside is delayed, and both the delayed and the referencechannel are input to the system. The software leads theuser thru the measurement, allowing calibration on aknown sideband, or inputting of the discriminatorconstant Ka.

Using the HP 3047A in the frequency discriminatormode allows the data collection and translation to be doneautomatically. The HP 3047A contains both a microwavephase detector and an RF phase detector.

ADVANTAGES OFHP 3047A SYSTEM USED IN

FREOUENCY DISCRI M I NATOR]tIETHOD

o Automotic colibrotion, conversionof units

o Hord-copy outputr Sensitivity follows spectro of

free-running sourceo Only one source required

95

DEMONSTRATION

Delay LinelMixerFrequency Discriminator Method

Using the HP 30474 System

34' Coble

The HP lI729B can also be used in the frequencydiscriminator mode. A single, fixed frequency 640 MHzsignal is still required. The DUT is downconverted, andthe discriminator placed at the IF. The necessary powersplitter and phase detector are already provided. The onlyexternal piece ofhardware needed is a simple length ofcable, which can be as inexpensive as a piece ofRG 55cable, since the highest operating frequency is less than1.3 GHz. The HP 86624 with internal FM modulation canbe used as a convenient calibration signal.

TYPICAL MEASUREMENT RESULTS

a

-Lg

-?g

-30

-a6

-59

-64

-?5

-ag

-95

- l @

- l l 9

- t?g

- l 3 g

- t a a

- l 5 g

- l 5 a

- 1 7 Il4 lAgK

! ( f ) t d B c / H z l v r , t H z l4k

HP 8656A at 640 MHzDelay Llneltllxer

Frequency Dlscrlmlnator MethodUslng the HP 30474 SYstem

FREOUENCY DISCRIMINATOR METHOD-IMPLETUIENTATION HP 117298'8662A

BosebondSpectrumAnolyzer

5 to 1280 MHz

o Discriminote ot HP117298 lF

96

wpfcAL sENstTtvtw oF Hp 117298t8662AIN FREOUENCY DISCRIMINATOH METHOD

MEE-RUNNIilG*:.ill:.:isYmEstrn

nt t:-or'

mo AT, o a . _

l0 MHz

l i_

Tlpical achievable system sensitivity is shown here. A34 ft.Jength ofcoaxial cable was used as the delayelement. Of course, longer delay lines would yieldincreased sensitivity, but reduced offset frequency range.

DEMONSTRATION

<10 MHz Noise Soeclrum

HP 8662A

Delay UneltllxerFrequency Dbcrlmlnator Method

Urlng the HP 11729818662A

TYPICAL MEASUREMENT RESUTTSGUNN OSC IA.sGHz !17294 FREO. OISC. XETHOO - OELAY-5SNS

START A HzRES BI 3AS Hz VAf lA Hz

STOP lsa kHzStP l9O ccc

Delay LlnelillxcrFrequency Dlscrlmlnator tethod

Uslng the HP 11729818662A

97

CONVERT TO Tff' IF DESIRED

3"+(l)c/11 = Lot \'/ ilL lo kHz loo kHz

X0 = nowured noisolsuaj - (dbm)- dotwtr,d (dDm)

calibrotion lavol- sot sidobondlo (d0c)

urrisr rotioDondw'idth Normolizotion - Gg)s/A c.oRnEclo N (it oppticobto) Ub)

- Lotoq tnftul Gb)x (t) = - 6\cltz) -

The HP 11729818662,4' in the frequency discriminatormode provide an easy solution for measurements onunstable or free-running sources. It can be used tomeasure sources to 18 GHz, yet the discriminator alwaysoperates less then 1.3 GHz. The one disadvantage of HP11729B,18662A implementation over a true'single source'technique is that the measurement is limited to the noisefloor of the multiplied 640 MHz signal.

HP 117298t8662A lNFREOUENCY DISCRIMINATOR METHOD

ADVANTAGESo Increosed sens i t i v i t y con be

obto ined w i th long l inewi thout p rob lem o f loss

. H P 8 6 6 2 4 c o n b e u s e dfor co l ib ro t ion

o B u i l t i n q u o d r o t u r e m o n i t o ro Broodbond d isc r im ino toro De loy l ine o f lF less expens ive

t o i m p l e m e n t t h o n

de loy l ine o f mic rowove

DISADVANTAGESo L imi ted to no ise f loor

o f mu l t ip l ied 640 MHz. Quodro ture se t by vory ing DUT

f requency or leng th o f l ine

FREOUENCY DISCRIMINATOR METHOD

ADVANTAGESr Requi res only one sourceo Low broodbond noise f looro Sensit ivity motches f ree

running VCO chorocter is t ico AM Suppress ion

x30 dB to ' 1 .5 GHzx15 dB > 1 .5 GHz

DISADVANTAGESo Poor close-in sensit ivityo Moy be d i f f icu l t to implement

deloy- l ine d iscr iminotorof microwove due toloss of l ine ond microphonics

o Other sensi t ive implemen-totions of microwoveore usuol ly norrowbond

-,In summary, the primary advantages of the frequencydiscriminator method are shown here. In practicefit maybe difficult to implement the delay line diicriminator at-microwave frequencies, as longer delay will improve thesensitivity but eventually the loss in the delay line willexceed the source power available and cancel any furtherimprovement. Also longer delay lines limit the maximumoffset frequency that can be measured.

5. Summary of Source Measurement lbchniques

SUM]UIARY

S o n o w I k n o w o l lPhose No ise .But wh ich method

T h n l r i e n a n r { e l

obout

do I use?

99

CHOICE OF METHOD DEPENDSON WHAT YOU WANT TO MEASUREI

FREE-FUNNING':: il i::"SYNTHESIZER

AT 10 GHz

i

o

f

d

3o

ao

I

ORO A1I GHz

10 MHzrTAL

The hrst criterion for deciding on a particular method oOphase noise measurement is how low the source or devicenoise is that is to be measured. Here again is acomparison ofthe sensitivities ofthe four phase noisemeasurement methods. Direct spectrum method is easiestif the source is stable and the noise is fairly high; timedomain has limited usefulness. Frequency discriminatorsensitivity follows the spectra offree-running sources; thephase detector method can have the lowest overallserusitiuity, depending on the reference source used.

Tb determine usable sensitivity in the phase detectormethod, the reference source must be specifred. Here arelow noise sources usable for references, from 10 MHz to 10 GHz.

PHASE DETECTOR METHODNOISE FLOOR WITH

TYPICAL REFERENCES

HP 86738 al l0 GHz

100

MEASURING A PHASE-LOCKEDMICROWAVE SOURCE

. locked to 100MHz input

oGood long-term stobil ity

,A" t l00H r rH I0 ̂ hz 00 . " .

0FtstT Fn0il ctinttlltd

Given what you know about the performance andcomplexity of the measurement methods, which methodwould you recommend to test the phase noise of a phase-locked microwave source with this level of expected phasenoise?

Which technique would you choose for this Gunn diodeVCO?WhY?

MEASURING A FREE-RUNNINGlilcRowAvE souRcE

o X - B o n d s o u r c ewi th tun ing

101

TIEASURING A HIGH-OUALITYLOW FREOUENCY OSCILLATOR

o High stobil i ty10 MHz crystol

r Good long-termstobility(<ppmldoy)

GENERAL SYSTEM CONSIDERATIONSor

how to make things easier with careful set-upo Heot sinking of sourceo Microphonic isolotion of sourceo Reduce frequency pulling effects

on source with circulotoro Choose coble to reduce microphonicsoThe RFI of the environmento "Cleon" power supplieso Resonont frequency of building!?

What about this high quality quartz oscillator?

With a better understanding of the fundamentals andmathematics of phase noise measurement, don't forgetthat the real world is still out there! Experience inmaking phase noise measurement quickly teaches thatcareful set-up and system environment are critical to goodmeasurements.

t02

Phase Noise MeasurementSystems Comparison

HP 117298Phase NoiseTest System

: HP 8566A/8 iHP 8568A/8

HP 3585AHP 3582AHP 3561A

!'jT 3l"Jy:".j'.,i

HP 3047APhase NoiseMeasurement

System

HP 3585ASpectrum Analyzer--

HP 3582A

HP 35601AS.A. lntertace

HP 3047A/117298prroPhase NoiseMeasuremenl

System

.005 - 18 GHz

yes

-120 clBc ai 10 kHz-132 dBc noise f loor

(w i th 117298)

r - - - - - - lI HP 86624 |I HP 8663A I, HP 8640AI HP 8672A I, HP 86738t Relerence sources IL - - - - - J

Principle olOperal ion

Frequency Range

Low Noise MicrowaveRelerence Signal

System NoiseSpec at 10 GHz

DUTo-B_rReference

Noise Spectrum(ottset lrequency)

<10 MHz (<40 MHz)

Two-port Noise yes

Delay Line Mode

AM-Noise

Manual Operation

Automatic Operation(HP-rB)

Sollware exlensive

'speci l icat ion shown requires a second, user supplied source of comparable phase norse.

yes

yes

HP 35854Speclrum Analyzer

HP 117298Phase Noise

Tesl Set

DUTo--e_rReference

DUT

o--b_rComparable

- '120 dBc al 10 kHz-132 dBc noise t loor

-160 dBc a t 10 kHz --160 dBc noise l loor

HP 117298Phase Noise

Test Set

HP 8662A orHP 8663A

Synthesized SigGer

This compares some of the performance parameters of HP's two-source phase noise measurement systems.

III. Measurement of Tlvo Port Phase Noise

Ttrrning to phase noise measurements on devices, ourselection of measurement method is significantly easier.Again, the phase noise added or contributed by a device iscalled two-port, or residual, or additive noise'

SETINAR AGENDA

l . Bosis of Phose Noise

Why is Phose Noise imPortont?Whot is Phose Noise?Whot couses Phose Noise?

Quont i fy ing Phose Noise

l l . Meosurement Techniqueson Sources

1. Direct Spectrum Method2. Heterodyne/Counter Method

3. Phose Detector Method4. FrequencY Discriminotor

Method5. Summory of Source

Meosurement Techniques

4 fff. Meosurement of Two-PortPhose Noise (Devices)

lV. Phose Noise Meosurementon Pulsed Corriers

V. AM Noise Meosurement

TWO-PORT NOISE MEASUREMENT(Slngle Frequency Slgnal Processor)

Spectrum Anolyzer

TWo-port phase noise characterization is done with amoilifi ecl frequency discriminator method. Actually, it'sdone exactly like the noise floor test for the frequencydiscriminator or phase detector method. A commonreference source is split and applied to two channelssimultaneously. Since the goal is to have the noise of thesource common to both channels and arrive correlated atthe phase detector, the delay in the two-paths is held asclose to the same as possible. The signals are set inquadrature with a phase shifter, and the-noisec-ontribution of the DUT can be measured.

L04

TWO-PORT NOISE MEASUREI'ENTCONCERNS'TIMITATIONS

1. AM << p]'t2. Typicol AM rejection

: 30 dB to 1 .5 GHzs 15 dB > 1 .5 GHz

3 . No ise f l oo r some osphose detector method

4. T ime deloy in unknown poth5. F i l ter ing ef fects of unknown poth

Tlvo-port noise measurements are made on a widevariety of sources, with measurements on TWT's one ofthe most common, and measurements on SAWs gaining asSAW technology advances.

Some of the areas of caution for a two-portmeasurement are the same as for an absolutemeasurement. However, because of the nature of two-portnoise, some new concerns arise. Tlpically, two-port noiseof devices is much, much less than the absolute noise ofsources, and it does not increase linearly as a function ofcarrier frgquency (nor does it necessarily follow thetypical I-'and f{ relationships ofsources as the carrier isapproached). Also, the two-port AM noise of a devicemight be of the same order of magnitude as its phasenoise. This means even more attention must be paid tosystem noise floor. For example, typically a good qualitysource is used as the reference, even though ideally allsource noise will be correlated out.

TYPICAL DUT'S WHERETWO-PORT NOISE IS MEASURED

TWT'S

SAW deloy l ines

ompl i f iers

mul t ip l iers

dividers

mixers

J TLru

E€-

a-_J

105

Ifthe output frequency ofthe DUT does not equal theinput frequency, then it is impossible to directly measurethe two-port noise with a single measurement.Commonly, a similar DUT (with assumed similar noise) isplaced in the reference path as well. The rms sum of noiseofboth devices is then measured. As in the phase detectormethod when using a similar source as a reference, themeasurement sets an upper limit on the noise of eachdevice. At each offset frequency, one ofthe devices is atleast 3 dB better. And if three devices with similar noisecan be measured, the noise ofone ofthe devices can becalculated.

Another alternative to measuring the two-port noise ofa frequency translating device is to use a synthesizer ineach path, instead of a second DUT. The limitation here isthat the absolute noise ofthe synthesizers must be belowthe two-port noise of the DUT. This might be workablesometimes at RF frequencies. However, for microwavefrequencies, the absolute noise ofsources is typicallymuch higher than the two-port noise of devices.

TWO.PORT NOISE TIEASURE]TENTFREQUENCY TRANSLATING DEVICE

Alternotive No. 1

Analyzer

o Meosure rms sum o f no iseof bo th dev ices

TWO.PORT NOISE IIEASUREMENTFREOUENCY TRANSLATING DEVICE

Alternotive No. 2

e Absolute noise of referenceMUST be below two-por tnoise of DUT

106

MAKING TWO.PORT PHASE NOISEMEASUREMENTS WITH THE

HP 30474

DE]TONSTRATION

Two-Port MeasurementUsing the HP 3047A System

shift

For two-port phase noise measurements, HP has reallyonly one solution - the HP 30474 Phase NoiseMeasurement System. The HP 3047A has been widelyand successfully used for two-port noise measurements, asits high level mixers yield a low system noise floor. (TheHP 117298 can not be used for two-port microwave phasenoise measurements because the microwave mixer is notdc coupled on the IF, nor is there access to both ports ofthemixer.)

TYPICAL MEASUREMENT RESULTS

z-tg-ea-39

-aa-st

-69-70-aa

-s9-Lg6- r l 9- t ? 9- l 3 s- l t a- r 5 9- r 5 g-t7a

I ! ( f ) t d B c / H z l " "

+ t H . l r s K

Two-Port Phase Nolse MeasurementUslng the HP 3047A System

107

IV. Phase Noise Measurements on Pulsed Carriers

PULSED PHASENOISE MEASUREMENTS

t c + P R F l +

o Offsets < 1/2 PRF

l +

Making phase noise measurements on pulsed sourcesbrings up a new set of complications. First, it is importantto recognize that, since the noise spectrum is repeated oneach spectral line, phase noise measurements are onlyvalid to less than 7z the PRF. (If the first PRF line wasequal to the carrier in amplitude, a phase noisemeasurement atYzthe PRF would yield noise data 3 dBtoo high.)

Frequency discriminators clo not lend themselves easilyto pulsed phase noise measurements, though some workhas been successfully done (eg, by F. Labarr at TRW). ForHP's hardware implementations, pulsed phase noisemeasurements can be done with some limitations withthe phase detector method. In general, it is necessary toinsert a LPF to remove the PRF feedthrough from feedingthru to the LNA, baseband analyzer, or phase lock loop.Also, because the test signal is now only present part ofthe time, it is usually necessary to increase the gain inthe phase lock loop to maintain quadrature. Finally, sincethe reference is on all the time (but the DUT pulsed), thesystem noise floor increases as a function ofduty cycle.

PHASE NOISE MEASURETIENTSON PUTSED SOURCES

o Insert Low Poss Filter (LPF)o lncreose goino Noise f loor increoses inversely

proport ionol to duty cycle

108

PULSED PHASE NOISEMEASUREMENTS ON SOURCES

wrTH HP 11729818662A13047A

a

O

o

Select K6, externol mixerCol ibrote on pulsed s ignolDuty cycles down to= 20%

It is possible to make phase noise measurements onpulsed sources with the HP 117298/8662A/3047A. Inorder to use the standard units without internalmodifications, an external phase detector is used at theHP 117298 IF output. The L port drive is provided by thefront panel signal of the HP 8662A. After the phasedetector, a low pass filter removes the sum mixingproducts and the PRF lines, and the resulting signal isapplied to the HP 3047A Signal Input port. Thisconfiguration allows measurement on pulsed sources withduty cycles down to about 207o.

Measured output from an HP 11740S system (HP11729818662N3047A). A 10 GHz source, 400 kHz PRF,withSUVo duty cycle was measured with a 250 kHz LPFinserted to remove the PRF. The measured values on thepulsed source are indistinguishable from the measuredvalues when operating the source in CW.

TYPICAT PERFORMANCE OFPUTSED HP 11729818662A13047A

- l t e

109

PULSED PHASE NOISE MEASUREMENTSwlTH HP 11729818662A

HP 8562A OPr.003

o Externol LPF switched ino Duty cycles down to (5%

o Disobles out-of-phose lock indicotoro Rising noise floor ond decreosing

offset freouencies os ofunction of duty cycle

For measurements on pulsed sources with lower dutycycles, the HP LI729B|8662A can be used. It is necessaryto slightly reconfigure the HP lI729B so that the LPF canbe placed before the low noise amplifier and the phaselock loop. (There are ports at the rear panel ofthe HPlL729B for the LPF; however, internal cable reroutingmust be done to switch the LPF into the sig:nal path.)Also, for very low duty cycles, a small external voltagemust be applied to the'Auxiliary Noise Spectrrrm Output'to adjust the balance ofthe diodes in the phase detector.With this configuration, phase noise measurements onpulsed sources with duty cycles down to <1Vo have beensuccessfully made.

Again, it's important to remember that noise floorincreases as a function ofduty cycle. Also, at very lowduty cycles, eventually the noise floor of components inthe HP Ll729B will dominate (in particular, the noise ofthe IF amplifier).

For lower duty cycles, there has been experimentationwith Sample and Hold circuitry, and a modified delay linetechnique. However, these are not implemented in HPinstrumentation.WHAT ABOUT

LOWER DUTY CYCLES?

o S o m p l e o n d h o l do Modif ied deloy l ine

technique (TRW)

110

PUTSED 'TWO-PORT" PHASE NOISEMEASURE]UIENTS WITH HP 3047A

o Switch out LNAo Use high power inputso Colibrote on pulsed signolo Duty cycles down

to s 10%

The HP 3047A can also be used for pulsed two-portphase noise measurements. Tlpically, it is necessary toswitch out the internal HP 3047A low noise amplifrer. Ifcalibration is done on a pulsed signal, no corrections tothe measured data needs to be made. Successfulmeasurements have been made to duty cycles less than20Vo.

V. AM Noise Measurement

ATII NOISE IIIEASUREIIENTS

fn(f) =

Spectrol Density ofone AM modulotion Sidebond

Totol Signol Power

AM NOISE vs. PHASE NOISE

Corresponding to the J (0 definition,qn'({) is defined asthe noise power in one AM modulation sideband, dividedby the total signal power, in units of dBc/Hz.

Though AM noise usually measured on signals, inpractice phase noise is more important (both technicallyand economically) than AM noise for three reasons.

1. The majority of high capacity (ie, expensive)communication systems use angle modulation.

2. AM noise in complex signal sources is usually<<phase noise.

3. The FM noise or phase noise on the Local Oscillatoris directly transferred to the input signal, whereasthe effects of the LO AM noise may be greatlymitigated by the use of balanced mixers.

rt2

There are two common methods of AM noisemeasurement; the first is the corollary to the phasedetector method of phase noise measurement. Here, adouble-balanced mixer is operated 180 degrees out ofphase (out-of-quadrature condition), to yield maximumsensitivity to AM fluctuations and minimum sensitivitvto phase fluctuations.

AilI NOISE MEASURETIENTS

Method #2 - Using An AM Detector

The second method (the corollary ofthe direct spectrumtechnique) simply measures the AM fluctuations with anAM detector. and then looks at the baseband fluctuations.

AM NOISE MEASUREMENTS

Method #1 - Using a Double-BalancedMixer as an AM Detector

:

// E^\lgJOut-ol-Ouad,gtureMonito.

Vpeak - - -

BasebandAnalyzer

v(r) 0

113

Both techniques have similar procedures, and are fairlystraightforward.

Direct AM noise measurements can be made with theHP 117298 Option 130. The HP 8662A with known AMsidebands is a convenient calibration source.

A]UINOISE MEASURET'ENT PROCEDURE

1. Col ibroteon known level s idebond/corr ierrot io , in out -of -quodroturecondi t ion

2. Meosure Vnoise

3. Corrections

+

Vnoise (dBm)V c o t ( d B m )s idebond/cor r ie r co lATTN.(used in method f1 )1 0 l o g B nS/A effects

dBc/Hz

Yn(f) : _T

+

AM NOISE MEASUREMENTS

5 MHz to'18 GHz

"Bench-Top' ryrlem wlth HP 117298 Opt. 13018662A

1) Col ibrote wi th HP 86624 w/Rv Sideoonds

2) Meosure

BosebondSpectrumAnolyzerHP 117298 Oot 130

The HP L1729B Option 130 can also be usedautomatically with the HP 3047A software. If properlycalibrated, the system software for phase noisemeasurements can also be used for AM noisemeasurements.

AM NOISE MEASUREMENTS

< 1 0 M H zNoise

Spectrum

2''

5 MHz to1 8 G H z

Autonetlc lyrtom ulth HP tt729B Opt t30r060!tArg0f?A1) Select "Phose Noise w/o vo l toge contro l , ,2) Col ibrote wi th HP 86624 w/AM s idebonds3) Meosure

SignolInput

HP 3047A

HP 117298 Oo t 130

DElUIONSTRATION

t-,

--

-

--1

*tta-|-

-l

-

ftE l"=y*'"TJ

SYMBOL

to

t

t m

fc

f ( t ) , v ( t )

A f ( t ) r 6 f ( t ) , 6 v ( t )

A o ( t ) r 6 e ( t )

td

Ka

Kd

s ( f )5

sa f ( f )

sy(f )

L ( f )

res Ftl

M ( f )

vupeak

vssb

Vs

5 rmsv

6

Jn

o 2 ( r )v

T

UNITS

Hz

Hz

Hz

Hz

Hz

Hz

rad

sec

VoIts,/rad

VoIts/Hz

rad2/Hz

Hz2t\z

- 1H z -

dBc/Hz

Hz

dBc/Hz

Volt,s

VoIt,

Volt,

vo|Es2/Hz

( rad)

GLOsSARY OF SYMBOLS

DESCRIPTION

Camier frequency

Fourier f requenry (sideband, offset,baseband)

Modulating frequency

Cornen frequency

Instantaneous frequency

Instantaneous frequency fluctuations

Instantaneous phase fluctuations

Delay tirne

Phase detector constant

Frequency di scriminator constant

Spectral density of phase fluctuations

Spectral density of frequencyf luctuat ions

Orre-sided spectral densityfractional frequenry fluctuations

Single-sideband signal to carr ier rat io

Residual FM

Spectral density of At.l noise

Peak voltage of sinusoidal beat signal

Single sideband voltage

Signal voltage

Pouer spectral density of voltagef luctuat ions

Modulation index

Bessel coeff ic ient

Al lan var iance

Period of sample

116

Nunber of tirne domain averages

Lf/f.^ over interval r longov

Np

k

F

G

B

Bn

a)

L8w

LHR

Watts

Joulesl(eIvin

Hz

Hz

radlsec

Hz

Hz

Noise power

Boltzman's constant

Noise Figure

Gain

Banduidth

Noise bandwidth

Angular velocity

Loop Bandwidth

Loop Holding Range

1 . 3 8 X 1 0 - 2 3

tt?

References

Hewlett-Packard Application Notes

pN11729B-i phase Noise Characterization of Microwave Oscillators - Phase Detector Method

AN 246-2 Measuring Phase Noise with the HP 3585A Spectrum Analyzer

AN 207 Understanding and Measuring Phase Noise in the Frequency Domain

AN 150-4 Spectrum Analysis. . . . . Noise Measurements

AN 283-1 Applications and Measurements of Low Phase Noise Signals Using the HP 8662A SynthesizedSignal Generator

AN Z7O-2 Automatic Phase Noise Measurements with the HP 85684 Spectrum Analyzer

Other References

Scherer, D., "Generation of Low Phase Noise Microwave Signals", HP RF and Microwave Measurement

Symposium, May, 1981

Scherer, D., "The'Art'of Phase Noise Measurements", HP RF and Microwave Measurement Symposium'

May,1983

Scherer, D., "Design Principles and Test Methods for Low Phase Noise RF and Microwave Sources", HP RF

and Microwave Measurement Symposium, October' 1978

Temple, R., ',Choosing a Phase Noise Measurement Technique - Concepts and lmplementations", HP RF and

Microwave Measurement Symposium, February, 1983

Fischer, M., "An Overview of Modern Techniques for Measuring Spectral Purity", Microwaves, Vol. 18' No' Zpp.66-75, July 1979

Fischer, M. "Frequency Stability Measurement Procedures", Proceedings, Eighth Annual PTTI Meeting,

Goddard Spac'e ftignt Center, Code 250, Greenbelt, Maryland, 1976' pp' 575-618

Howe, D., ',Frequency Domain Stability Measurements: A Tutorial Introduction", N.B.S. Technical Note 679'

National Bureau of Standards, Boulder' CO' 1976

Shoaf, J.H., Halford, D., Risley, A.S., "Frequency Stability Qpecification and Measurement: High Frequency

and'Microwave Signals, t{.e.S. fecnnicdt ttot6 632, Natiirnal Bureau of Standards, Boulder, CO' 1973

Ashley, R.J. et al, "Measurement of Noigein Microwave Transmitters", IEEE Transactions on Microwave

Theory and Techniques, Vol. Mfi'%, No. 4, pp. 294-318, April,1977

Lance, Seal, Hudson, M6ndoza, and Halford, "Phase Noise Measurements Using Cross-_Spectrum Analysis"'

Conference on Precision Electromagnetic Measurements, Ottowa, Canada, June' 1978

Lance, Seal, Mendoza, Hudson, "Phase Noise Characteristics of Frequency Sources", Ninth Annual PTTI

Meeting, Goddard Space Flight Center, Greenbelt, Maryland, November, 1977

Lance, Seal, Labaar, "Phase Noise Measurement Systems"' TRW

Lance, Seal, "Phase Noise and AM Noise Measurements in the Frequency Domain at Millimeter Wave

Frequencies", TRW

Labaar, F., "A New Type of Detay Line FM Discriminator"' TRW

Ondria, F.G., 'A Microwave System for Measurements of AM and FM Noise Spectra", IEEE Transactions on

Microwave Theory and Techniques, Vol. MTT-16, pp.767-781, September' 1968

Robbins, W.P., "Phase Noise in Signal Sources", IEEE Telecommunications Series No. 9' 1982

118

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